Method and device for transmitting OFDM signal, and method and device for receiving OFDM signal

ABSTRACT

A method of transmitting, by a transmitting device, an orthogonal frequency division multiplexing (OFDM) signal in a wireless communication system, the method including: generating, by a digital module of the transmitting device, a frequency-shifted OFDM baseband signal by performing frequency up-shift of a first signal by a difference between a carrier frequency    0  and a first frequency    base , wherein the first frequency f base  is, among frequencies corresponding to integer multiples of 128Δ ; closest to the carrier frequency    0 , and wherein Δ  is an OFDM subcarrier spacing; up-converting, by an analog oscillator of the transmitting device, the frequency-shifted OFDM baseband signal by the first frequency    base  to generate an OFDM symbol signal at the carrier frequency fo; and transmitting the OFDM symbol signal at the carrier frequency    0 .

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a National Stage application under 35 U.S.C. § 371of International Application No. PCT/KR2018/015249, filed on Dec. 4,2018, which claims the benefit of Korean Application No.10-2018-0079450, filed on Jul. 9, 2018, Korean Application No.10-2018-0039089, filed on Apr. 4, 2018, U.S. Provisional Application No.62/629,714, filed on Feb. 13, 2018, and U.S. Provisional Application No.62/621,058, filed on Jan. 24, 2018. The disclosures of the priorapplications are incorporated by reference in their entirety.

TECHNICAL FIELD

The present disclosure relates to a wireless communication system. Moreparticularly, the present disclosure relates to a method and device fortransmitting an OFDM signal, and a method and device for receiving anOFDM signal.

BACKGROUND ART

In a mobile communication system, a transmitting device typicallygenerates a baseband signal, up-converts the baseband signal to a highercarrier frequency, and transmits an up-converted radio signal at thecarrier frequency. A receiving device then receives the radio signal,and down-converts the received radio signal from the carrier frequencyto a lower baseband frequency to obtain a baseband signal for furtherprocessing.

DISCLOSURE Technical Problem

If the information about the frequency for upconversion is unknown tothe transmitting device and the receiving device, mismatch may occurbetween the upconversion frequency used by the transmitting device andthe downconversion frequency used by the receiving device. The mismatchbetween the upconversion frequency and the downconversion frequencycauses a sudden phase change in each time symbol in the receivingdevice. Such a sudden phase change greatly degrades performance ofsignal recovery by channel estimation in the receiving device.Therefore, a method for reducing the phase change in each symbol causedby mismatch between the upconversion frequency and the downconversionfrequency, mismatch between the carrier frequency and the centerfrequency of the frequency band, or mismatch between the carrierfrequency and the center of the RF filter is required.

In addition, when the carrier frequency changes in the same frequencyband, a method of easily adjusting the carrier frequency without RFretuning is required.

Technical Solution

The object of the present disclosure can be achieved by techniquesdisclosed herein for transmitting an orthogonal frequency divisionmultiplexing (OFDM) signal by a transmitting device in a wirelesscommunication system. In one aspect, provided herein is a method oftransmitting, by a transmitting device, an orthogonal frequency divisionmultiplexing (OFDM) signal in a wireless communication system. Themethod comprises: generating, by a digital module of the device, afrequency-shifted OFDM baseband signal by performing frequency up-shiftof a first signal by a difference between a carrier frequency f₀ and afirst frequency f_(base); up-converting, by an analog oscillator of thedevice, the frequency-shifted OFDM baseband signal by the firstfrequency f_(base) to generate an OFDM symbol signal at the carrierfrequency f₀; and transmitting, by a transmitter of the device, the OFDMsymbol signal at the carrier frequency f₀. The first frequency f_(base)may be, among frequencies corresponding to integer multiples of 128Δf,closest to the carrier frequency f₀. Δf is an OFDM subcarrier spacing.

In another aspect, provided herein is a method of receiving, by a deviceat a receiving side, an orthogonal frequency division multiplexing(OFDM) signal in a wireless communication system. The method comprises:receiving an OFDM symbol signal at a carrier frequency f₀;down-converting, by an analog oscillator of the device, the OFDM symbolsignal by a first frequency f_(base) to generate a down-converted OFDMsymbol signal; and generating, by a digital module of the device, anOFDM baseband signal by performing frequency down-shift of thedown-converted OFDM symbol signal by a difference between the carrierfrequency f₀ and f_(base). The first frequency f_(base) may be, amongfrequencies corresponding to integer multiples of 128Δf, closest to thecarrier frequency f₀. Δf is an OFDM subcarrier spacing.

In a further aspect, provided herein is a device at a transmitting sidefor transmitting an orthogonal frequency division multiplexing (OFDM)signal in a wireless communication system. The device may comprise: adigital module; an analog oscillator; at least one antenna; at least oneprocessor; and at least one computer memory that is operably connectableto the at least one processor and that has stored thereon instructionswhich, when executed, cause the at least one processor to performoperations. The operations may comprise: generating, by the digitalmodule, a frequency-shifted OFDM baseband signal by performing frequencyup-shift of a first signal by a difference between a carrier frequencyf₀ and a first frequency f_(base): up-converting, by the analogoscillator, the frequency-shifted OFDM baseband signal by the firstfrequency f_(base) to generate an OFDM symbol signal at the carrierfrequency f₀; and transmitting, using the at least one antenna, the OFDMsymbol signal at the carrier frequency f₀. The first frequency f_(base)may be, among frequencies corresponding to integer multiples of 128Δf,closest to the carrier frequency f₀. Δf is an OFDM subcarrier spacing.

In a still further aspect, provided herein is a device at a receivingside for receiving an orthogonal frequency division multiplexing (OFDM)signal in a wireless communication system. The device may comprise: atleast one antenna; an analog oscillator; a digital module; at least oneprocessor; and at least one computer memory that is operably connectableto the at least one processor and that has stored thereon instructionswhich, when executed, cause the at least one processor to performoperations. The operations may comprise: receiving, using the at leastone antenna, an OFDM symbol signal at a carrier frequency f₀;down-converting, by the analog oscillator, the OFDM symbol signal by afirst frequency f_(base) to generate a down-converted OFDM symbolsignal; and generating, by the digital module, an OFDM baseband signalby performing frequency down-shift of the down-converted OFDM symbolsignal by a difference between the carrier frequency f₀ and f_(base).The first frequency f_(base) may be, among frequencies corresponding tointeger multiples of 128Δf, closest to the carrier frequency f₀. Δf isan OFDM subcarrier spacing.

In each aspect at the transmitting side, the digital module may beconfigured to implement an inverse fast Fourier transform (IFFT) on thefirst signal.

In each aspect of a transmitting side, performing the frequency up-shiftof the first signal by the difference between f₀ and f_(base) maycomprise: up-shifting, by N_(frac), a resource mapping for the firstsignal that is input to the IFFT, where N_(frac) is an integersatisfying f₀−f_(base)=N_(frac)*Δf.

In each aspect at the transmitting side, the digital module may comprisea digital oscillator. Performing the frequency up-shift of the firstsignal by the difference between f₀ and f_(base) may be performed by thedigital oscillator.

In each aspect at the transmitting side, the digital oscillator mayreset, before transmitting the OFDM symbol signal, a phase of the OFDMsymbol signal to a predetermined value at an end of a cyclic prefix ofthe OFDM symbol signal.

In each aspect at the receiving side, the digital module may beconfigured to implement a fast Fourier transformer (FFT) on thedown-converted OFDM symbol signal.

In each aspect at the receiving side, performing the frequencydown-shift of the down-converted OFDM symbol signal by the differencebetween f₀ and f_(base) may comprise: down-shifting, by N_(frac), aresource de-mapping from the FFT for the down-converted OFDM symbolsignal, where N_(frac) is an integer satisfying f₀−f_(base)=N_(frac)*Δf.

In each aspect at the receiving side, the digital module may comprise adigital oscillator. Performing the frequency down-shift of thedown-converted OFDM symbol signal by the difference between f₀ andf_(base) may be performed by the digital oscillator.

In each aspect at the receiving side, the digital oscillator may reset aphase of the down-converted OFDM symbol signal to a predetermined valueat an end of a cyclic prefix of the down-converted OFDM symbol signal.

The above technical solutions are merely some parts of theimplementations of the present disclosure and various implementationsinto which the technical features of the present disclosure areincorporated can be derived and understood by persons skilled in the artfrom the following detailed description of the present disclosure.

Advantageous Effects

According to the present invention, phase change according to symbolswhich occurs due to mismatch between the upconversion frequency and thedownconversion frequency may be easily minimized. Accordingly, even ifthe upconversion frequency is unknown to the transmitting device and thereceiving device, or mismatch between the upconversion/downconversionfrequency and the center of the RF filter or mismatch between thecarrier frequency and the center frequency of a cell occurs, the signalrecovery performance at the receiving device may be maintained.

In addition, when the carrier frequency changes in the same frequencyband, the carrier frequency may be easily adjusted without RF retuning.

DESCRIPTION OF DRAWINGS

FIGS. 1A and 1B illustrate examples of modulation and upconversion of abaseband signal to a carrier frequency;

FIGS. 2A and 2B are diagrams illustrating examples of phase changeaccording to a difference between an upconversion frequency and adownconversion frequency;

FIG. 3 illustrates an example of resetting the phase at a symbolboundary;

FIGS. 4A and 4B illustrate examples of generation of a baseband signaland modulation and upconversion to a carrier frequency thereof accordingto some implementations of the present disclosure;

FIGS. 5A to 5C are diagrams illustrating examples of Implementation 1 ofthe present disclosure;

FIGS. 6A and 6B are diagrams illustrating examples of Implementation 2-1of the present disclosure;

FIGS. 7A and 7B are diagrams illustrating examples of resource mappingaccording to Implementation 2-1 of the present disclosure and resourcemapping according to Implementation 2-2 of the present disclosure;

FIGS. 8A and 8B are diagrams illustrating examples of Implementation 2-2of the present disclosure;

FIGS. 9A to 9C are diagrams illustrating examples of Implementation 3 ofthe present disclosure;

FIGS. 10A and 10B are diagrams illustrating examples of Implementationa2-1 of the present disclosure;

FIGS. 11A and 11B are diagrams illustrating examples of Implementationa2-2 of the present disclosure;

FIG. 12 is a diagram illustrating another use example of the presentdisclosure;

FIGS. 13A and 13B illustrate examples of a transmitter structure and areceiver structure according to some implementations the presentdisclosure; and

FIG. 14 is a block diagram illustrating examples of components of atransmitting device and a receiving device according to someimplementations of the present disclosure.

MODE FOR INVENTION

Wireless communication systems typically communicate using specificranges of radio frequencies (RF). To ensure proper transmission in theseRF ranges, wireless systems typically implement, at the transmitter, atechnique called upconversion to convert signals from a lower frequencyrange to a higher (RF) frequency range, and also implement, at thereceiver, a technique called downconversion to convert signals from ahigher (RF) frequency range to a lower frequency range.

However, difficulties may arise where information about frequencyconversion is unknown to a transmitting device and/or a receivingdevice. In such scenarios, mismatch may occur between an upconversionfrequency used by the transmitting device and a downconversion frequencyused by the receiving device. Such mismatch between upconversion anddownconversion frequencies may cause a phase offset in each time symbolreceived at the receiving device. Phase offset may degrade performanceof signal recovery by channel estimation in the receiving device.

Furthermore, in some scenarios, mismatches may occur between a carrierfrequency and a center frequency of a frequency band, or between thecarrier frequency and a center of an RF filter. Such mismatches may alsoresult in phase offsets in the received time symbols, which may degradereception performance.

Therefore, difficulties may arise in systems where such phase offsetoccurs due to mismatch between upconversion and downconversionfrequencies, or due to mismatch between the carrier frequency and thecenter frequency of the frequency band, or due to mismatch between thecarrier frequency and the center of the RF filter. In addition, when thecarrier frequency changes in the same frequency band, difficulties mayarise in adjusting the carrier frequency without performing RF retuning.

Implementations disclosed herein enable a transmitter that is configuredto perform upconversion in a manner that mitigates or removes such phaseoffset. In some implementations, the transmitter upconverts frombaseband to one of a finite number of frequencies that arepredetermined, and thus configured to result in no phase offset at thereceiver. Since these finite number of frequencies may be different fromthe actual carrier frequency utilized by the transmitter, thetransmitter may compensate for any such difference by pre-shifting thebaseband signal by that difference.

In some implementations, the pre-shifting may be implemented either byperforming frequency domain shifting (e.g., shifting an input of aninverse Fast Fourier Transform (IFFT) at the transmitter) or may beimplemented by time-domain shifting (e.g., shifting an output of theIFFT, for example, using a digital oscillator).

Analogously, in some implementations, a receiver is configured toperform downconversion from one of the finite number of predeterminedfrequencies down to baseband. Again, since the finite number offrequencies may be different from the actual carrier frequency utilizedby the receiver, the receiver may compensate for any such difference bypost-shifting the resulting baseband signal by that difference.

Accordingly, implementations disclosed herein may mitigate or removephase offsets that occur due to mismatch between the upconversionfrequency and the downconversion frequency. Thus, even if theupconversion frequency is unknown to the transmitting device and thereceiving device, or even if mismatch between theupconversion/downconversion frequency and the center of the RF filteroccurs, or even if mismatch between the carrier frequency and the centerfrequency of a cell occurs, the signal recovery performance at thereceiving device may be maintained.

In addition, in some scenarios, when the carrier frequency changes inthe same frequency band, the carrier frequency may be easily adjustedwithout RF retuning.

Reference will now be made in detail to various implementations of thepresent disclosure, examples of which are illustrated in theaccompanying drawings. The detailed description, which will be givenbelow with reference to the accompanying drawings, is intended toexplain exemplary implementations of the present disclosure, rather thanto show the only implementations that can be implemented according tothe disclosure. The following detailed description includes specificdetails in order to provide a thorough understanding of the presentdisclosure. However, it will be apparent to those skilled in the artthat the present disclosure may be practiced without such specificdetails.

In some instances, known structures and devices are omitted or are shownin block diagram form, focusing on important features of the structuresand devices, so as not to obscure the concept of the present disclosure.The same reference numbers will be used throughout this specification torefer to the same or like parts.

The following techniques, apparatuses, and systems may be applied to avariety of wireless multiple access systems. Examples of the multipleaccess systems include a code division multiple access (CDMA) system, afrequency division multiple access (FDMA) system, a time divisionmultiple access (TDMA) system, an orthogonal frequency division multipleaccess (OFDMA) system, a single carrier frequency division multipleaccess (SC-FDMA) system, and a multicarrier frequency division multipleaccess (MC-FDMA) system. CDMA may be embodied through radio technologysuch as universal terrestrial radio access (UTRA) or CDMA2000. TDMA maybe embodied through radio technology such as global system for mobilecommunications (GSM), general packet radio service (GPRS), or enhanceddata rates for GSM evolution (EDGE). OFDMA may be embodied through radiotechnology such as institute of electrical and electronics engineers(IEEE) 802.11 (Wi-Fi), IEEE 802.16 (WiMAX), IEEE 802.20, or evolved UTRA(E-UTRA). UTRA is a part of a universal mobile telecommunications system(UMTS). 3rd generation partnership project (3GPP) long term evolution(LTE) is a part of evolved UMTS (E-UMTS) using E-UTRA. 3GPP LTE employsOFDMA in DL and SC-FDMA in UL. LTE-advanced (LTE-A) is an evolvedversion of 3GPP LTE. For convenience of description, implementations ofthe present disclosure are described herein as being applied to the 3GPPbased communication system, especially, the NR system. However, thetechnical features of the present disclosure are not limited thereto.Although the following detailed description is based on a mobilecommunication system corresponding to the 3GPP NR system, it isapplicable to any other mobile communication systems except uniquefeatures of 3GPP NR. For example, the present disclosure is applicableto a communication technology in which the upconversion frequency andthe downconversion frequency are not shared in advance between thetransmitting device and the receiving device and communicationtechnologies in which mismatch may occur between the upconversionfrequency and the center of the RF filter or between theupconversion/downconversion frequency and the center frequency of acell.

In the present disclosure, a user equipment (UE) may be a fixed ormobile device. Examples of a UE include various devices that transmitand receive user data and/or various kinds of control information to andfrom a base station (BS). The UE may be referred to as a terminalequipment (TE), a mobile station (MS), a mobile terminal (MT), a userterminal (UT), a subscriber station (SS), a wireless device, a personaldigital assistant (PDA), a wireless modem, a handheld device, etc. Inaddition, in the present disclosure, a base station (BS) generallyrefers to a fixed station that performs communication with a UE and/orwith another BS, and exchanges various kinds of data and controlinformation with the UE and/or another BS. The BS may be referred to asan advanced base station (ABS), a node-B (NB), an evolved node-B (eNB),a base transceiver system (BTS), an access point (AP), a processingserver (PS), etc. In particular, the base station of the UTRAN isreferred to as Node-B, the base station of E-UTRAN is referred to aseNB, and the base station of the new radio access technology network isreferred to as gNB.

In the present disclosure, a node refers to a fixed point configured totransmit/receive a radio signal through communication with a UE. Varioustypes of eNBs may be used as nodes irrespective of the terms thereof.For example, a BS, a node B (NB), an e-node B (eNB), a pico-cell eNB(PeNB), a home eNB (HeNB), a relay, a repeater, etc. may be a node. Inaddition, the node may not be a BS. For example, the node may be a radioremote head (RRH) or a radio remote unit (RRU). The RRH or RRU generallyhas a lower power level than a power level of a BS. Since the RRH or RRU(hereinafter, RRH/RRU) is generally connected to the BS through adedicated line such as an optical cable, cooperative communicationbetween RRH/RRU and the BS can be smoothly performed in comparison withcooperative communication between BSs connected by a radio line. Atleast one antenna may be installed per node. The antenna may be aphysical antenna or an antenna port or a virtual antenna.

In the present disclosure, a cell may refer to a prescribed geographicalarea to which one or more nodes provide a communication service.Accordingly, in the present disclosure, communicating with a specificcell may include communicating with a BS or a node which provides acommunication service to the specific cell. In addition, a DL/UL signalof a specific cell refers to a DL/UL signal from/to a BS or a node whichprovides a communication service to the specific cell. A node providingUL/DL communication services to a UE is called a serving node and a cellto which UL/DL communication services are provided by the serving nodeis especially called a serving cell.

A 3GPP-based communication system typically implements a cell in orderto manage radio resources, and a cell associated with the radioresources is distinguished from a cell of a geographic region. Forexample, a “cell” of a geographic region may be understood as a coveragewithin which a node can provide service using a carrier, and a “cell” ofa radio resource is associated with bandwidth (BW) which is a frequencyrange configured by the carrier. Since DL coverage, which is a rangewithin which the node is capable of transmitting a valid signal, and ULcoverage, which is a range within which the node is capable of receivingthe valid signal from the UE, depends upon a carrier carrying thesignal, the coverage of the node may be associated with coverage of the“cell” of a radio resource used by the node. Accordingly, the term“cell” may be used to indicate service coverage of the node sometimes, aradio resource at other times, or a range that a signal using a radioresource can reach with valid strength at other times. The “cell”associated with the radio resources is defined by combination ofdownlink resources and uplink resources, that is, combination of DLcomponent carrier (CC) and UL CC. The cell may be configured by downlinkresources only, or may be configured by downlink resources and uplinkresources.

For terms and technologies not specifically described among the termsand technologies used in this specification, 3GPP LTE/LTE-A standarddocuments such as 3GPP TS 36.211, 3GPP TS 36.212, 3GPP TS 36.213, 3GPPTS 36.321 and 3GPP TS 36.331 and 3GPP NR standard documents such as 3GPPTS 38.211, 3GPP TS 38.212, 3GPP 38.213, 3GPP 38.214, 3GPP 38.215, 3GPPTS 38.321, 3GPP TS 38.300 and 3GPP TS 38.331 can be referenced.

Referring to the standard 3GPP TS 36.211, for all physical signals andphysical channels except the physical random access channel, OFDM symbolbaseband signals, e.g., single carrier frequency division multipleaccess (SC-FDMA) baseband signals, are generated as follows. In the LTEsystem, the time-continuous signal s_(l)(t) in SC-FDMA symbol l in anuplink slot is defined for the time interval 0≤t<(N_(CP,l)+N)∝T_(s)(where the fast Fourier transform (FFT) size N is equal to 2048) by thefollowing equation.

$\begin{matrix}{{s_{l}(t)} = {\overset{{\lceil{N_{RB}^{UL}{N_{sc}^{RB}/2}}\rceil} - 1}{\sum\limits_{k = {- {\lfloor{N_{RB}^{UL}{N_{sc}^{RB}/2}}\rfloor}}}}{a_{k^{( - )},l} \cdot e^{j\; 2\;{\pi{({k + {1/2}})}}\Delta\;{f{({t - {N_{{CP},l}T_{s}}})}}}}}} & {{Equation}\mspace{14mu} 1}\end{matrix}$

where k⁽⁻⁾=k+└N_(RB) ^(UL)N_(sc) ^(RB)/2┘, the subcarrier spacing isΔf=15 kHz, and a_(k,l) is the content of resource element (k,l). Theindex k is an index numbered from 0 to N^(UL) _(RB)×N^(RB) _(sc)−1 inthe frequency domain, and l is an index numbered from 0 to N^(UL)_(symb)−1 in the time domain.

In an LTE system, the uplink transmitted signal in each slot utilizes aresource grid of N^(UL) _(RB)×N^(RB) _(sc) subcarriers and N^(UL)_(symb) OFDM symbols. Each resource element in the resource grid isuniquely defined by the index pair (k,l) in a slot, where k=0, . . . ,N^(UL) _(RB)×N^(RB) _(sc)−1 and l=0, . . . , N^(UL) _(symb)−1. Theexpression N^(UL) _(RB) denotes the number of resource blocks (RBs) inan uplink (UL) slot and depends on the uplink transmission bandwidthconfigured in a cell. The expression N^(RB) _(sc) denotes the number ofsubcarriers constituting one RB. In the LTE system, N^(RB) _(sc)=12. TheRB is defined as 12 consecutive subcarriers in the frequency domain. Theexpression T_(s) is a basic time unit for LTE, whereinT_(s)=1/(15*10³*2048) seconds. For reference, the sampling time isdefined as 1/(N_(FFT)*Δf), where N_(FFT) is the FFT size (equal to theIFFT size) and Δf is the subcarrier spacing. When N_(FFT)=2048 and thebasic subcarrier spacing is Δf=15 kHz, the basic time unit T_(s) of theLTE system corresponds to the sampling time. The expression N^(UL)_(symb) denotes the number of SC-FDMA symbols in the UL slot, whereinN^(UL) _(symb)=7 for the normal cyclic prefix (CP) and N^(UL) _(symb)=6for the extended CP. The expression N_(CP,l) is the cyclic prefixlength. The following table lists the values of N_(CP,l) used on anuplink of the LTE system.

TABLE 1 Cyclic prefix Configuration length N_(CP,l) Normal cyclic 160for l = 0 prefix 144 for l = 1,2,...,6 Extended cyclic 512 for l =0,1,...,5 prefix

The SC-FDMA symbols in a slot are transmitted in increasing order of l,starting with l=0, where SC-FDMA symbol l>0 starts at a time, within theslot, given by the expression

$\sum\limits_{l^{\prime} = 0}^{l - 1}{( {N_{{CP},l^{\prime}} + N} ){T_{s}.}}$

The time-continuous signal s_(l) ^((p))(t) on antenna port p inorthogonal frequency division multiplexing (OFDM) symbol l in a downlinkslot is defined for 0≤t<(N_(CP,l)+N)∝T_(s) by the following equation.

$\begin{matrix}{{s_{l}^{(p)}(t)} = {{\sum\limits_{k = {- {\lfloor{N_{RB}^{DL}{N_{sc}^{RB}/2}}\rfloor}}}^{- 1}{a_{k^{( - )},l}^{(p)} \cdot e^{j\; 2\;\pi\; k\;\Delta\;{f{({t - {N_{{CP},l}T_{s}}})}}}}} + {\sum\limits_{k = 1}^{\lceil{N_{RB}^{DL}{N_{sc}^{RB}/2}}\rceil}{a_{k^{( + )},l}^{(p)} \cdot e^{j\; 2\;\pi\; k\;\Delta\;{f{({t - {N_{{CP},l}T_{s}}})}}}}}}} & {{Equation}\mspace{14mu} 2}\end{matrix}$

where k⁽⁻⁾=k+└N_(RB) ^(DL)N_(sc) ^(RB)/2┘, and k⁽⁺⁾=k+└N_(RB)^(DL)N_(sc) ^(RB)/2┘−1. In the time interval 0≤t<(N_(CP,l)+N)∝T_(s), thevariable N equals 2048 for subcarrier spacing Δf=15 kHz and equals 4096for subcarrier spacing Δf=7.5 kHz. The OFDM symbols in a slot aretransmitted in increasing order of l, starting with l=0, where OFDMsymbol l>0 starts at time

$\sum\limits_{l^{\prime} = 0}^{l - 1}{( {N_{{CP},l^{\prime}} + N} )T_{s}}$within the slot. The index k is an index numbered from 0 to N^(DL)_(RB)×N^(RB) _(sc)−1 in the frequency domain, and l is an index assignedvalues from 0 to N^(DL) _(symb)−1 in the time domain.

In the LTE system, the downlink transmitted signal in each slot isdescribed by a resource grid of N^(DL) _(RB)×N^(RB) _(sc) subcarriersand N^(DL) _(symb) OFDM symbols. Each resource element in the resourcegrind is uniquely identified by the index pair (k,l) in a slot, wherek=0, . . . , N^(DL) _(RB)×N^(RB) _(sc)−1 and l=0, . . . , N^(UL)_(symb)−1. N^(DL) _(RB) denotes the number of RBs in a DL slot anddepends on the downlink transmission bandwidth configured in a cell.N^(DL) _(symb) denotes the number of OFDM symbols in the DL slot,wherein N^(DL) _(symb)=7 for the normal cyclic prefix (CP) and N^(DL)_(symb)=6 for the extended CP. N_(CP,l) is the cyclic prefix length. Thefollowing table lists the values of N_(CP,l) used on downlink in the LTEsystem.

TABLE 2 Cyclic prefix Config- length uration N_(CP,l) Normal Δƒ = 15 kHz160 for l = 0 cyclic 144 for l = prefix 1,2,...,6 Extended Δƒ = 15 kHz512 for l = cyclic 0,1,...,5 prefix Δƒ = 7.51 kHz 1024 for l = 0,1,2

FIGS. 1A and 1B illustrate examples of modulation and upconversion of abaseband signal to a carrier frequency that may be implemented in somesystems (e.g., an LTE system). In particular, FIG. 1A illustrates anexample of modulation and upconversion, to the carrier frequency, of acomplex-valued SC-FDMA baseband signal for each antenna port, and FIG.1B illustrates an example of modulation and upconversion, to the carrierfrequency, of the complex-valued OFDM baseband signal for each antennaport.

Filtering may be performed prior to uplink transmission, for example asspecified by the standard 3GPP TS 36.101, and likewise filtering may beperformed prior to downlink transmission, for example as specified bythe standard 3GPP TS 36.104. In the examples of FIGS. 1A and 1B, thefrequency f₀ is the upconversion frequency. In some scenarios (e.g., inan LTE system), the upconversion frequency may be set equal to a carrierfrequency of a cell.

In some systems (e.g., in an LTE system), since a synchronization signalof the cell and a physical broadcast channel (PBCH) of the cell aretransmitted in a total of six RBs around the carrier frequency of thecell, the user equipment (UE) of the LTE system can know the downlinkcarrier frequency of the cell by acquiring the synchronization signaland the PBCH. In such scenarios, if the UE and the base station (BS)know the downlink carrier frequency, then they may also know the uplinkcarrier frequency in scenarios where (i) the downlink carrier frequencyand the uplink carrier frequency are the same, for example, in the caseof time division duplex (TDD), or where (ii) the uplink carrierfrequency used together with the downlink carrier frequency ispredetermined, for example, in the case of frequency division duplex(FDD), or where (iii) the uplink carrier frequency is explicitlybroadcast through the system information of the cell or the like. As aresult, in such scenarios of the LTE system, both the UE and the basestation (BS) may know the carrier frequency of a cell used fortransmission/reception of a radio signal.

In a legacy LTE system, the following frequencies are configured to bethe same: (i) the center of a radio frequency (RF) filter (e.g., afilter between IFFT and upconversion, a filter applied afterupconversion, etc.), (ii) the center frequency of the carrier bandwidth,and (iii) the upconversion frequency f₀. In addition, the same frequencyis used for up-converting the baseband to the carrier frequency signaland for down-converting a radio signal to the baseband signal.

However, with the increase in various utilizations of Machine TypeCommunication (MTC), Internet of things (IoT) communications, andultra-reliable and low latency communication (URLLC), a new radio accesstechnology (NR) different from the legacy LTE communication technologyis under development. The NR system considers use of frequencies abovethe frequency band used in the legacy communication system and alsoconsiders supporting a bandwidth much wider than the frequency bandwidthused in the legacy communication system. Considering the drawbacks ofthe legacy LTE system, in which it is difficult to introduce acommunication technology having forward compatibility due to variousconstraints, the NR system is being developed so as to reduce suchconstraints and thereby facilitate introduction of future communicationtechnology having forward compatibility with the NR system.

Accordingly, in the NR system, the frequency used for upconversion ofthe baseband signal is not necessarily limited to the center frequencyof the cell. In addition, in the NR system, the frequency resourcethrough which the synchronization signal is transmitted is notnecessarily limited to be the center of the frequency band of the cell.

Considering that the UE may not be able to support a wide bandwidth tobe supported in the NR system at one time, the UE may be configured tooperate in a part of the frequency bandwidth (hereinafter, bandwidthpart (BWP)) of the cell. The BWP may be assigned based on any referencepoint. The reference point is not necessarily limited to be the centerfrequency of the cell. If only a part of the frequency bandwidth of thecell is used for communication such as BWP-based communication andNB-IoT, the receiving device may not know the upconversion frequencyused by the transmitting device before downconversion of the receivingsignal.

Accordingly, the upconversion frequency for the baseband signal may bedifferent from the downconversion frequency for the radio signal, andthe upconversion frequency is not necessarily limited to be the centerof the RF filter.

In addition, it is expected that a variety of numerologies will besupported in the NR system. If the numerology for the same frequencyband changes, the subcarrier spacing may change. This change insubcarrier spacing may result in change in the upconversion frequency orthe downconversion frequency. Therefore, there is a need for a techniqueby which the transmitting device and the receiving device can easilyadjust the upconversion frequency and the downconversion frequency,respectively.

Before explaining implementations of the present disclosure in furtherdetail, the basic frame structure and physical resources of the NRsystem so far discussed will be described in order to facilitateunderstanding of the present disclosure.

In the description of the present disclosure, unless otherwise noted,the size of various fields in the time domain is expressed in time unitsT_(c)=1/(Δf_(max)*N_(f)), where Δf_(max)=480*10³ Hz and N_(f)=4096, orin time units T_(s). T_(c) is the basic time unit for NR. The constantκ=T_(s)/T_(c)=64, where T_(s)=1/(Δf_(ref)*N_(f,ref)), Δf_(ref)=15*10³Hz, N_(f,ref)=2048. T_(s) is the basic time unit for LTE. In NR,multiple OFDM numerologies are supported as given by the followingtable, where μ and the cyclic prefix for a bandwidth part are given bythe higher-layer parameters provided by a BS.

TABLE 3 μ Δƒ = 2^(μ)* 15 [kHz] Cyclic prefix 0 15 Normal 1 30 Normal 260 Normal, Extended 3 120 Normal 4 240 Normal

Downlink and uplink transmissions are organized into frames withT_(f)=(Δf_(max)N_(f)/100)*T_(c)=10 ms duration, each consisting ofT_(sf)=(Δf_(max)N_(f)/1000)*T_(c)=1 ms duration. The number ofconsecutive OFDM symbols per subframe N_(symb) ^(subframe,μ)=N_(symb)^(slot)N_(slot) ^(subframe,μ). Each frame is divided into twoequally-sized half-frames of five subframes. There is one set of framesin the uplink and one set of frames in the downlink on a carrier.

For subcarrier spacing configuration μ, slots are numbered n_(s)^(μ)∈{0, . . . , N_(slot) ^(subframe,μ)−1} in increasing order within asubframe. There are N_(symb) ^(slot) consecutive OFDM symbols in a slotwhere N_(symb) ^(slot) depends on the cyclic prefix as given by Table 4and Table 5. Table 4 shows the number of OFDM symbols per slot, thenumber of slots per frame, and the number of slots per subframe, for thenormal cyclic prefix, and Table 5 shows the number of OFDM symbols perslot, the number of slots per frame, and the number of slots persubframe, for the extended cyclic prefix.

TABLE 4 μ N^(slot) _(symb) N^(frame, μ) _(slot) N^(subframe, μ) _(slot)0 14 10 1 1 14 20 2 2 14 40 4 3 14 80 8 4 14 160 16 5 14 320 32

TABLE 5 μ N^(slot) _(symb) N^(frame, μ) _(slot) N^(subframe, μ) _(slot)2 12 40 4

In Table 4 and Table 5, N_(symb) ^(slot) denotes the number of symbolsper slot, N_(slot) ^(frame,μ) is the number of slots per frame forsubcarrier configuration μ, N_(slot) ^(subframe,μ) is the number ofslots per subframe for subcarrier configuration μ.

For each numerology and carrier, a resource grid of N_(grid,x)^(size,μ)N_(sc) ^(RB) subcarriers and N_(symb) ^(subframe,μ) OFDMsymbols is defined, starting a common resource block N_(grid) ^(start,μ)indicated by higher-layer signaling by a BS, where N_(grid,x) ^(size,μ)is the size of the resource grid and N_(sc) ^(RB) is the number ofsubcarriers per resource block. There is one set of resource grids pertransmission direction (DL or UL) with the subscript x set to DL and UL.The subscript x is DL for downlink and UL for uplink. When there is norisk for confusion, the subscript x may be dropped. There is oneresource grid for a given antenna port p, subcarrier spacingconfiguration μ, and transmission direction (downlink or uplink). Eachelement in the resource grid for antenna port p and subcarrier spacingconfiguration p is called a resource element and is uniquely identifiedby (k,l)_(p,μ) where k is the index in the frequency domain and l refersto the symbol position in the time domain relative to some referencepoint. Resource element (k,l)_(p,μ) corresponds to the complex valuea_(k,l) ^((p,μ)). When there is no risk for confusion, or no particularantenna port or subcarrier spacing is specified, the indices p and μ maybe dropped, resulting in a_(k,l) ^((p)) or a_(k,l).

A resource block (RB) is defined as N^(RB) _(sc)=12 consecutivesubcarriers in the frequency domain. Reference resource blocks arenumbered from 0 and upward in the frequency domain. Subcarrier 0 ofreference resource block 0 is common for all subcarrier spacingconfigurations μ, also denoted as ‘reference point A’ or ‘point A’, andserves as a common reference point for other resource block grids.Reference point A is obtained from the higher-layer parameter providedby a BS. Common resource blocks (CRBs) are numbered from 0 and upwardsin the frequency domain for subcarrier spacing configuration μ.Subcarrier 0 of common resource block 0 for subcarrier spacingconfiguration μ coincides with the reference point A. The relationbetween the common resource block number n_(CRB) in the frequency domainand resource elements (k, l) for subcarrier spacing configuration μ isgiven by the following equation.

$\begin{matrix}{n_{CRB}^{\mu} = \lfloor \frac{k}{N_{sc}^{RB}} \rfloor} & {{Equation}\mspace{14mu} 3}\end{matrix}$where k is defined relative to subcarrier 0 of the resource grid forsubcarrier spacing configuration μ.

In the NR system, physical resource blocks (PRBs) are defined within acarrier bandwidth part and numbered from 0 to N_(BWP,i) ^(size)−1, wherei is the number of the carrier bandwidth part and N_(BWP,i) ^(size) isthe size of bandwidth part i. The relation between physical and commonresource blocks in carrier bandwidth part i is given by the followingequation.n _(CRB) =n _(PRB) +N _(BWP,i) ^(start)  Equation 4

where N_(BWP,i) ^(start) is the common resource block where the carrierbandwidth part starts relative to common resource block 0.

A bandwidth part is a subset of contiguous common resource blocksdefined for a given numerology μ_(i) in the bandwidth part i on a givencarrier. The starting position N_(BWP,i) ^(start,μ) and the number ofresource blocks N_(BWP,i) ^(size,μ) in a bandwidth part shall fulfillN_(grid,x) ^(start,μ)≤N_(BWP,i) ^(start,μ)<N_(grid,x)^(start,μ)+N_(grid,x) ^(size,μ) and N_(grid,x) ^(start,μ)<N_(BWP,i)^(size,μ)+N_(BWP,i) ^(start,μ)+N_(grid,x) ^(size,μ). A UE can beconfigured with the certain number (e.g. up to four) of bandwidth partsin the downlink with a single downlink bandwidth part being active at agiven time. A UE can be configured with the certain number (e.g. up tofour) of bandwidth parts in the uplink with a single uplink bandwidthpart being active at a given time.

In some wireless communication systems, the carrier frequency used bythe transmitter and the receiver are known to each other, and thetransmitter and the receiver set the same carrier frequency as theupconversion frequency and the downconversion frequency, respectively.However, due to inaccuracy of the analog oscillator or the phase-lockedloop (PLL), an error, i.e., a frequency offset, occurs between thefrequencies generated by the transmitter and the receiver. In this case,the signal phase varies depending on the symbols at the receiving end.However, generally, phase change due to inaccuracy of the analog moduleis not so serious as to make channel estimation with the referencesignal (RS) useless and, generally, such phase change does not greatlyaffect the received signal recovery.

On the other hand, in a radio communication system such as an NR systemsupporting a wideband cell, the UE and the BS may have to operatewithout the information on a carrier frequency for upconversion known tothe UE and the BS. Therefore, when the UE and the BS performupconversion and downconversion using different carrier frequencies, thephase of the receiving device may change abruptly in each symbol as willbe described later even if it is assumed that there is no frequencyoffset, i.e., frequency error, resulting from the inaccuracy of theanalog oscillator or the PLL.

The time-continuous signal s_(l) ^((p,μ))(t) on antenna port p andsubcarrier spacing configuration μ for OFDM symbol l in a subframe forany physical channel or signal except the physical random access channel(PRACH) is defined by the following equation.

$\begin{matrix}{{s_{l}^{({p,\mu})}(t)} = {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot e^{j\; 2\;{\pi{({k + k_{0}^{\mu} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - {N_{{CP},l}^{\mu}T_{c}}})}}}}}} & {{Equation}\mspace{14mu} 5}\end{matrix}$

where 0≤t<(N_(u) ^(μ)+N_(CP,l) ^(μ))T_(c). Equation 5 may be expressedas:

$\begin{matrix}{{s_{l}^{({p,\mu})}(t)} = {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot e^{j\; 2\;{\pi{({k + k_{0}^{\mu} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - {N_{{CP},l}^{\mu}T_{c}} - t_{{start},l}^{\mu}})}}}}}} & {{Equation}\mspace{14mu} 6}\end{matrix}$

where t_(start,l) ^(μ)≤t<t_(start,l) ^(μ)+(N_(u) ^(μ)+N_(CP,l)^(μ))T_(c) is the time within the subframe.

In Equation 5 and Equation 6, the value of k₀ ^(μ) is obtained from thehigher-layer parameter k0 provide by a BS, and is such that the lowestnumbered subcarrier in a common resource block for subcarrier spacingconfiguration μ coincides with the lowest numbered subcarrier in acommon resource block for any subcarrier spacing configuration less thanμ. The starting position t_(start,l) ^(μ) of OFDM symbol l forsubcarrier spacing configuration μ in a subframe is given as follows.

$\begin{matrix}{t_{{start},l}^{\mu} = \{ \begin{matrix}0 & {l = 0} \\{t_{{start},{l - 1}}^{\mu} + {( {N_{u}^{\mu} + N_{{CP},{l - 1}}^{\mu}} ) \cdot T_{c}}} & {otherwise}\end{matrix} } & {{Equation}\mspace{14mu} 7}\end{matrix}$

Here, the effective symbol length N^(μ) _(u) of OFDM symbol l and thecyclic prefix (CP) length N^(μ) _(CP,l) of OFDM symbol l are given as:

$\begin{matrix}\begin{matrix}{N_{u}^{\mu} = {2048{\kappa \cdot 2^{- \mu}}}} \\{N_{{CP},l}^{\mu} = \{ \begin{matrix}{512{\kappa \cdot 2^{- \mu}}} & {{extended}\mspace{14mu}{cyclic}\mspace{14mu}{prefix}} \\{{144{\kappa \cdot 2^{- \mu}}} + {16\kappa}} & {{{normal}\mspace{14mu}{cyclic}\mspace{14mu}{prefix}},} \\\; & {l = {0\mspace{14mu}{or}\mspace{14mu}{7 \cdot 2^{\mu}}}} \\{144{\kappa \cdot 2^{- \mu}}} & {{{normal}\mspace{14mu}{cyclic}\mspace{14mu}{prefix}},} \\\; & {1 \neq {0\mspace{14mu}{and}\mspace{14mu} l} \neq {7 \cdot 2^{\mu}}}\end{matrix} }\end{matrix} & {{Equation}\mspace{14mu} 8}\end{matrix}$

The time continuous signal s_(l) ^((p,μ))(t) on antenna port p for PRACHis defined by the following equation.

$\begin{matrix}{{{s_{l}^{({p,\mu})}(t)} = {\sum\limits_{k = 0}^{L_{RA} - 1}{a_{k}^{({p,{RA}})} \cdot e^{j\; 2\mspace{11mu}{\pi{({k + {Kk}_{0} + \overset{\_}{k}})}}\Delta\;{f_{RA}{({t - {N_{{CP},l}^{RA}T_{c}}})}}}}}}{k = {\Delta\;{f/\Delta}\; f_{RA}}}} & {{Equation}\mspace{14mu} 9}\end{matrix}$

where 0≤t<(N_(u)+N_(CP,l) ^(RA))T_(c). A detailed description of eachparameter in Equation 9 can be found in 3GPP TS 38.211.

The transmitting device up-converts the OFDM symbol baseband signals_(l) ^((p,μ))(t) for the antenna port p and the subcarrier spacingconfiguration μ to the uplink frequency f_(Tx) using the free-runningoscillator of the frequency f_(Tx). The upconversion of the OFDM symbolbaseband signal s_(l) ^((p,μ))(t) for the antenna port p and thesubcarrier spacing configuration μ to the upconversion frequency f_(Tx)may be expressed as:

$\begin{matrix}{{x^{({p,\mu})}(t)} = {{{s_{l}^{({p,\mu})}(t)} \cdot e^{j\; 2\;\pi\; f_{Tx}t}} = {\sum\limits_{k = 0}^{{N_{RB}^{\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot e^{j\; 2{\pi{({k + k_{0} - {N_{RB}^{\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot e^{j\; 2\pi\; f_{Tx}t}}}}} & {{Equation}\mspace{14mu} 10}\end{matrix}$

In Equation 10, N_(RB) ^(μ) may be the number of RBs for the subcarrierspacing configuration μ. N_(RB) ^(μ) may be N_(grid,x) ^(size,μ).N_(grid,x) ^(size,μ) is a value configured by the BS, and the UE mayknow N_(grid,x) ^(size,μ) through the system information. Since a signalthat is actually transmitted in a final signal that the transmittingdevice obtains by multiplying a transmitted signal by e^(j2πf) ^(Tx)^(t) for the frequency upconversion (modulation) is a real signal ratherthan a complex signal, the real value of the final signal of Equation 10is transmitted. That is, modulation and upconversion of thecomplex-valued OFDM symbol baseband signal for the antenna port p andthe subcarrier spacing configuration μ to the upconversion frequencyf_(Tx) may be expressed as follows.x ^((p,μ))(t)=Re{s _(l) ^((p,μ))(t)·e ^(j2πf) ^(Tx) ^(t)}  Equation 11

Even if the transmitting device transmits only the real value of thecomplex signal, the receiving device applies FFT after converting thereceived signal back into the complex signal. Therefore, in thedescription of the present disclosure, the transmitted signal isexpressed as a complex signal for convenience, and is equivalent to thereal signal in modeling. The same applies to the reception operation.

When a radio signal x^((p,μ))(t) is received by the receiving device,the receiving device performs frequency downconversion on x^((p,μ))(t)to obtain the baseband signal {circumflex over (x)}^((p,μ))(t). When itis assumed that the receiving device uses an arbitrary frequency f_(Rx)in frequency downconversion, the frequency downconversion of the radiosignal x^((p,μ))(t) for the antenna port p and the subcarrier spacingconfiguration μ may be expressed as:

$\begin{matrix}{{{\hat{x}}^{({p,\mu})}(t)} = {{{x^{({p,\mu})}(t)} \cdot e^{{- j}\; 2\;\pi\; f_{Rx}t}} = {\sum\limits_{k = 0}^{{N_{RB}^{\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot e^{j\; 2{\pi{({k + k_{0} - {N_{RB}^{\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot e^{j\; 2\pi\;{({f_{Tx} - f_{Rx}})}t}}}}} & {{Equation}\mspace{14mu} 12}\end{matrix}$

To show the phase change of the received signal {circumflex over(x)}^((p,μ))(t), the frequencies f_(Tx) and f_(Rx) may be expressed asf_(TX)=N_(Tx)*Δf+Δ_(offset) and f_(Rx)=N_(Rx)*Δf+Δ_(offset) where theterm Δf is the subcarrier spacing, the term N_(Tx) is a positive integerclosest to f_(Tx)/Δf (e.g., floor{f_(Tx)/Δf} or ceil{f_(Tx)/Δf}), theterm N_(Rx) is a positive integer closest to f_(Rx)/Δf (e.g.,floor{f_(Rx)/Δf} or ceil{f_(Rx)/ΔP}), and the term Δ_(offset) is a realnumber whose magnitude is smaller than Δf. In the description of thepresent disclosure, f_(Tx) and f_(Rx), are expressed using the sameΔ_(offset) for simplicity, but Δ_(offset) may differ between f_(Tx) andf_(Rx).

Using these expressions, Equation 12 may be rearranged as given below.

$\begin{matrix}\begin{matrix}{{{\hat{x}}^{({p,\mu})}(t)} = {{x^{({p,\mu})}(t)} \cdot e^{{- j}\; 2\pi\; f_{Rx}t}}} \\{= {{{s_{l}^{({p,\mu})}(t)} \cdot e^{{{j2}\;\pi\; f_{Tx}t}\;} \cdot e^{{- j}\; 2\;\pi\; f_{Rx}t}} \simeq {{s_{l}^{({p,\mu})}(t)} \cdot e^{j\; 2\;{\pi{({f_{Tx} - f_{Rx}})}}t}}}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{e^{j\; 2{\pi{({k + k_{0} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} \\{e^{j\; 2{\pi{({f_{Tx} - f_{Rx}})}}t}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ {e^{j\; 2{\pi{({k + k_{0} - {N_{grid}^{{size}.\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - t_{{start}.l} - {N_{{CP}.l}^{\mu}T_{c}}})}}} \cdot} } \\{e^{{{j\; 2{\pi\Delta}\;{{f{({N_{Tx} - N_{Rx}})}} \cdot {({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}}\}} \cdot}} \\{e^{j\; 2{\pi\Delta}\;{{f{({N_{Tx} - N_{Rx}})}} \cdot {({t - t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}}}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size}.\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ {e^{j\; 2{\pi{({k + k_{0} - {N_{grid}^{{size}.\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - t_{{start}.l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} } \\{ e^{j\; 2{\pi\Delta}\;{{f{({N_{Tx} - N_{Rx}})}} \cdot {({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \} \cdot e^{j\;{\psi l}}}\end{matrix} & {{Equation}\mspace{14mu} 13}\end{matrix}$

Even in an environment in which there is no frequency offset, which is afrequency error unintentionally produced by the characteristics of thetransmitter/receiver components, the received signal {circumflex over(x)}^((p,μ))(t) may suffer phase change byΨ_(l)=2πΔf(N_(Tx)−N_(Rx))·(t_(start,l)+N_(CP,l) ^(μ)T_(c)) in frequencyupconversion or frequency downconversion if f_(Tx) is not equal tof_(Rx) for the following reason. In Equation 5, if expressing t−N_(CP,l)^(μ)T_(c) as t′, t−N_(CP,l) ^(μ)T_(c) in s_(l) ^((p,μ))(t), i.e., timet′ at which inverse fast Fourier transform (IFFT) is applied is definedonly for T_(CP)≤t′<T_(OFDM) (i.e., N_(CP,l) ^(μ)T_(c)≤t′<N_(u)^(μ)T_(c)), but t in e^(j2πf) ^(Tx) ^(t), which is a frequencyupconversion component, namely, the upconversion time t for which thefree-running oscillator operates is defined as −∞<t<∞.

FIGS. 2A and 2B are diagrams illustrating examples of phase changeaccording to a difference between an upconversion frequency and adownconversion frequency in terms of a device and a signal waveform.

Referring to FIG. 2A, an information symbol a_(k) that transmittingdevice intends to transmit is converted to an OFDM baseband signal s(t)through IFFT. The transmitting device up-converts s(t) to s(t)·e^(j2πf)^(Tx) ^(t) using a free-running oscillator of frequency f_(Tx). Whens(t)·e^(j2πf) ^(Tx) ^(t) arrives at the receiving device over a radiochannel, if signal distortion in the radio channel is not taken intoconsideration, the receiving device down-converts s(t) e^(j2πf) ^(Tx)^(t) to s′(t) by multiplying s(t)·e^(j2πf) ^(Tx) ^(t) by e^(j2πf) ^(Rx)^(t) using the free-running oscillator (OSC) with the frequency f_(Rx),and performs FFT on s′(t), thereby obtaining an information symbola′_(k).

Referring to FIG. 2B, an OFDM symbol signal is obtained when a cyclicprefix (CP) is added to the IFFT signal obtained by performing IFFT onthe information symbol. The CP added to the IFFT signal causes atransition with respect to the waveform of the IFFT signal in the timedomain. As a result, when the OFDM symbol signal is loaded on the signalof the free-running OSC, the phase of the transmitted signal may not bezero at the beginning of the OFDM symbol. In addition, the phase of thetransmit/receive signal may be different among the beginnings of theOFDM symbols.

Therefore, if f_(Tx) is not equal to f_(Rx), performance issignificantly degraded in the signal recovery process through channelestimation at the receiving end due to the abrupt phase change betweenthe symbols caused by the difference between f_(Tx) and f_(Rx). If thephase is abruptly changed between the OFDM symbols, the receiver cannotapply a channel estimation value obtained using the reference signal(RS) of a specific OFDM symbol to other OFDM symbols, or the receivedsignal may not be restored properly when the channel estimation value isused. It is not appropriate for the transmitter to insert an RS in everyOFDM symbol to allow the receiver to correctly estimate the channelstate of each symbol because the RS overhead becomes excessively large.

Several types of techniques may be utilized for an NR system to mitigatesuch issues of phase discontinuity/mismatch between symbols. Someexamples of such techniques are described below, along with potentialdisadvantages of each.

-   -   Technique A: The gNB informs the UE of the carrier frequency        that the gNB uses, and the UE compensates for the corresponding        phase discontinuity.

According to this scheme, if the BS transmits a transmitted signalwithout separately performing pre-compensation thereon, the UE performscompensation for phase discontinuity for each symbol using the carrierfrequency information of the BS. For example, the UE, which is thereceiver, performs phase compensation so as to cancel the phasediscontinuity occurring due to e^(j2π(f) ^(Tx) ^(−f) ^(Rx) ^()t) inEquation 12 on a symbol-by-symbol basis. In addition, when the UEtransmits a signal, the UE serving as a transmitting side performspre-compensation for the phase discontinuity term, and the BS performsreception on the assumption that the carrier frequencies of the BS andthe UE are equal to each other. However, this technique may bedisadvantageous in that both the BS and the UE must implement two modessince operations of the BS and the UE before the information about thecarrier frequency used by the BS is transmitted have to be additionallydefined as well as operations of the BS and the UE after the informationabout the carrier frequency used by the BS is transmitted.

-   -   Technique B: The BS, which is a transmitter, performs phase        pre-compensation using the DL carrier frequency information of        the UE.

This technique may be implemented, for example, in the NB-IOT system asan operation before the receiver receives the information about thecarrier frequency in Implementation A. For example, the transmitterperforms phase pre-compensation to cancel the phase discontinuityoccurring due to e^(j2π(f) ^(Tx) ^(−f) ^(Rx) ^()t) in Equation 12 on asymbol-by-symbol basis. In this case, the receiver only needs to operateassuming that the carrier frequencies of the transmitter and thereceiver coincide with each other. However, in this technique, when thebandwidth part having different frequency positions for UEs isconfigured as in the NR system, the BS must perform phasepre-compensation using different values for each UE. Accordingly, withthis technique, the receiver operation of the UE becomes very simple,but the transmitter operation of the BS becomes very complicated.

-   -   Technique C: the transmitter and the receiver perform phase        pre-compensation assuming a common reference point.

In this technique, the transmitter does not use the information aboutthe carrier frequency of the receiver (and the receiver does not use theinformation about the carrier frequency of the transmitter). Instead, acommon reference point is predefined between the transmitter and thereceiver and phase pre-compensation for the reference point is performedon a symbol-by-symbol basis. For example, the transmitter performs phase(pre-)compensation on the phase discontinuity occurring due to e^(j2π(f)^(Tx) ^(−f) ^(commmon) ^()t), and the receiver performs phase(pre-)compensation on the phase discontinuity occurring due to e^(j2π(f)^(common) ^(−f) ^(Rx) ^()t). As a particular example, in some scenariosEquation 5, above, may be modified for phase pre-compensation asfollows.

$\begin{matrix}{{s_{l}^{({p,\mu})}(t)} = {\lambda_{l}^{\mu}{\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot e^{j\; 2{\pi{({k + k_{0}^{\mu} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - {N_{{CP},l}^{\mu}T_{c}}})}}}}}}} & {{Equation}\mspace{14mu} 14}\end{matrix}$

Here, Δf_(ref)=15 kHz, and

λ_(l)^(μ) = e^(−j 2 π ⋅ (p_(μ)Δ f_(ref) + Δ)(N_(CP, l)^(μ)T_(C) + t_(start, l)^(μ))),where

$P_{\mu} - {{\min( {\underset{k \in Z^{+}}{\arg\mspace{14mu}\min}{{f_{0} - {M \times 5\; k\;{Hz}} - {k\;\Delta\; f_{ref}}}}} )}.}$Here M={−1, 0, 1} for band between 0 to 2.65 GHz and M=0 for the otherbands. Δ, the phase compensation value, is determined between thequantized carrier frequency and the non-quantized carrier frequency,where Δ=0 for the quantized carrier frequency, Δ=f₀−p_(μ)Δf_(ref) forthe non-quantized carrier frequency. Here f₀ is the carrier frequency ofthe receiver and k is a variable. Therefore, k that minimizes theabsolute value of ‘f₀−M*5 kHz−kΔf_(ref)’ may be p_(μ). This technique,however, is disadvantageous in that phase compensation is alwaysperformed by both the transmitter and the receiver. In addition,according to this technique, the transmitter and the receiver calculatethe phase for each symbol based on the carrier frequency thereof andapply the compensation term to the signal. Therefore, assuming that allavailable frequencies, that is, all frequencies to which subcarriers canbe mapped, can become carrier frequencies, the phase compensation termbecomes a function of a very high resolution and a very longperiodicity, requiring a very complex implementation.

For reference, the NR standard does not explicitly specify a particulartechnique for implementation. In the NR standard, a modulation andupconversion technique is defined as shown in the following table belowsuch that the transmitting side and the receiving side respectivelyreset the carrier frequency to zero phase on the symbol-by-symbol basisto maintain a certain value of the phase of the carrier frequency at thestarting point of each symbol (see 3GPP TS 38.211 section 5.4). This isspecified in the standard document 3GPP TS 38.211 V15.1.0 as follows.

TABLE 6 5.4 Modulation and upconversion Modulation and upconversion tothe carrier frequency ƒ₀ of the complex-valued OFDM baseband signal forantenna port p, subcarrier spacing configuration μ, and OFDM symbol l ina subframe assumed to start at t = 0 is given by Re{s_(l)^((p, μ))(t)·e^(j2πƒ) ₀ ^((t−t) ^(μ) ^(start,l) ^(−N) ^(μ) ^(CP,l) ^(T)^(c) ⁾} for all channels and signals except PRACH and by Re{s_(l)^((p,μ))(t)·e^(j2πƒ) ⁰ ^(t)} for PRACH.

FIG. 3 illustrates an example of resetting the phase at a symbolboundary. That is, FIG. 3 is a diagram illustrating an example of phasecompensation defined in the NR standard. In FIG. 3, T_(CP) correspondsto t_(start,l) ^(μ)N_(CP,l) ^(μ)T_(c) of Table 6. If the carrierfrequency used by the transmitting side to transmit a signal does notmatch the carrier frequency used by the receiving side to receive thesignal, the signals down-converted at the receiving side will havedifferent phases according to symbols. Referring to Table 6 and FIG. 3,in the frequency upconversion process, a time shift is performed for thetransmitted signal by t_(start,l) ^(μ)+N_(CP,l) ^(μ)T_(c) to reset thephase. Therefore, the phase discontinuity occurring in each symbol dueto the carrier frequency is eliminated at the transmitting side and thereceiving side, and consequently, phase discontinuity/mismatch betweensymbols is eliminated from the signal received by the receiving side.This may be expressed as the following equation.

$\begin{matrix}{{{\hat{x}}^{({p,\mu})}(t)} = {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot e^{j\; 2{\pi{({k + k_{0} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot e^{j\; 2{{\pi{({f_{Tx} - f_{Rx}})}} \cdot {({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}}}}} & {{Equation}\mspace{14mu} 15}\end{matrix}$

When Equation 15 is rearranged to explain how Equation 15 appears in anactual implementation, the follow equation is obtained.

$\begin{matrix}\begin{matrix}{{{\hat{x}}^{({p{,\mu}})}(t)} = {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ {e^{j\; 2{\pi{({k + k_{0} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} } \\{ e^{j\; 2{{\pi{({f_{Tx} - f_{Rx}})}} \cdot {({{- t_{{start},l}} - {N_{{CP},l}^{\mu}T_{c}}})}}} \} \cdot e^{j\; 2\;{\pi{({f_{Tx} - f_{Rx}})}}t}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ {e^{j\; 2{\pi{({k + k_{0} - {N_{grid}^{{size}.\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} } \\{ {e^{{- j}\; 2{\pi\Delta}\;{f_{Tx} \cdot {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})} \cdot}}e^{{j2}\;\pi\; f_{Tx}t}} \} \cdot} \\{e^{j\; 2\pi\;{f_{Rx} \cdot {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot e^{{- j}\; 2\;\pi\; f_{Rx}t}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size}.\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ {e^{j\; 2{\pi{({k + k_{0} - {N_{grid}^{{size}.\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - t_{{start}.l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} } \\{ {e^{j\;\psi_{{Tx},l}} \cdot e^{{j2}\;\pi\; f_{Tx}t}} \} \cdot e^{j\;\psi_{{Rx},l}} \cdot e^{{- 2}\;\pi\; f_{Rx}t}}\end{matrix} & {{Equation}\mspace{14mu} 16}\end{matrix}$

Techniques for adding phase discontinuity/mismatch described thus fardefine a phase reset at the carrier frequency level. For actualupconversion (or downconversion) of the carrier frequency level,components such as a phase-locked loop (PLL) and a mixer are used.

FIGS. 4A and 4B are examples of generation of a baseband signal andmodulation and upconversion to a carrier frequency thereof, according tosome implementations of the present disclosure.

Referring to FIGS. 4A and 4B, for example, a carrier frequency foractual upconversion (or downconversion) is generated using the PLL, andthe mixer or the like is used for upconversion to the carrier frequency.Components such as the PLL and the mixer are implemented as analogdevices or operate at very high speeds, and therefore in some scenariosit may be difficult to implement a phase reset at the carrier frequencylevel at the transmitting and receiving sides.

In other words, referring to Table 6, the NR standard specifies that aphase reset should be implemented by directly controlling the phase ofthe carrier frequency. However, in some scenarios, it may be difficultin reality to directly control the phase of the carrier frequency.Therefore, due to this practical difficulty of directly controlling thephase at the carrier frequency, some systems perform upconversion anddownconversion using a carrier frequency having a continuous phase atthe carrier frequency level, and further implement, at the basebandlevel, a phase reset function for eliminating the resulting phasediscontinuity/mismatch between the symbols caused byupconversion/downcoversion at the analog level.

In such systems, referring to Equation 16, upconversion anddownconversion at the carrier frequency level using a carrier frequencyhaving a continuous phase corresponds to e^(j2πf) ^(Tx) ^(t) ande^(j2πf) ^(Rx) ^(t), respectively. The frequencies f_(Tx) and f_(Rx) maybe arbitrary frequencies to which subcarriers are mapped, and may beexpressed as f_(Tx)=N_(Tx)*Δf+Δ_(offset) andf_(Rx)=N_(Rx)*Δf+Δ_(offset), respectively, using the subcarrier spacingΔf. Herein, the subscripts Tx and Rx denote the transmitting side andthe receiving side, respectively. As described in regards to Equation12, above, the term N_(Tx) is a positive integer closest to f_(Tx)/Δf(e.g., floor{f_(Tx)/Δf} or ceil{f_(Tx)/Δf}), the term N_(Rx) is apositive integer closest to f_(Rx)/Δf (e.g., floor{f_(Rx)/Δf} orceil{f_(Rx)/Δf}), and the term Δ_(offset) is a real number whosemagnitude is smaller than Δf. In the description of the presentdisclosure, f_(Tx) and f_(Rx) are expressed using the same Δ_(offset)for simplicity, but Δ_(offset) may differ between f_(Tx) and f_(Rx).Referring to Equation 16, the phase reset function at the baseband levelcorresponds to e^(jΨ) ^(Tx,l) and e^(jΨ) ^(Rx,l) .

Therefore, in such systems, the transmitting side and the receiving sidecompensate the phase using only their respective carrier frequencies,which correspond to Ψ_(Tx,l)=−2πf_(Tx)(t_(start,l)+N_(CP,l) ^(μ)T_(c))and Ψ_(Rx,l)=2πf_(Rx)(t_(start,l)+N_(CP,l) ^(μ)T_(c)) in Equation 16,respectively.

This corresponds to the transmitting side performing phase compensationassuming that the receiver uses the direct current (DC) tone, i.e., 0,as the carrier frequency for downconversion, and the receiving sideperforming phase compensation assuming that the transmitter uses the DCtone as the carrier frequency for upconversion. In such scenarios, ifthe transmitting side and the receiving side operate without informationabout the carrier frequency, then the term e^(j2π(f) ^(Tx) ^(−f) ^(Rx)^()t) in Equation 13 is equal to e^(j2π(f) ^(Tx) ^(−0)t)·e^(j2π(0−f)^(Rx) ^()t).

Therefore, the current NR standard (3GPP TS 38.211 V15.1.0) may beunderstood as specifying that the transmitting side utilizes e^(j2πf)^(Tx) ^((t−T) ^(CP) ⁾ for upconversion on the assumption that thecarrier frequency used for downconversion is 0 and, analogously, thatthe receiving side utilizes e^(j2πf) ^(Rx) ^((t−T) ^(CP) ⁾ fordownconversion on the assumption that the carrier frequency used by thetransmitting side for upconversion is 0.

As discussed above (regarding Techniques A to C), some wirelesscommunication systems that are based on the NR standard may apply aphase compensation term by calculating the phase in every symbol basedon the carrier frequency. However, considering the myriad applicable oravailable frequencies as f_(XX) (where XX is Tx or Rx), this phasecompensation term may become a function of a very high resolution and avery long periodicity, potentially imposing a very compleximplementation.

To address such challenges, the present disclosure describesimplementations for addressing scenarios where the transmitter and thereceiver operate without knowing the carrier frequency used fortransmission or without knowledge of the upconversion/downconversionfrequency.

Techniques for Lowering Complexity

The current NR standard defines the following numerologies (see, e.g.,section 5.4 of 3GPP TS 38.101-1 and section 5.4 of 3GPP TS 38.101-2).

TABLE 7 Frequency Range Range of range covered ΔF_(Global) F_(REF-Offs)N_(REF-Offs) N_(REF) FR1 0 ~ 3000  5 kHz 0 MHz 0 0 ~ MHz 599999 FR1 3000~ 15 kHz 30000 600000 600000 ~ 24250 MHz 2016666 MHz FR2 24250 ~ 60 kHz24250.8 2016667 2016667 ~ 100000 MHz 3279165 MHz

TABLE 8 Frequency Range SS Block frequency Range of range coveredposition SS_(REF) GSCN GSCN FR1 0 ~ 2700 N * 900 kHz + M* 3N + 1 ~ MHz[TBD 70 ~ 100 kHz] M − 1 [8999] N = 1:3000, M = −1:1 FR1 2700 ~ 2400MHz + N * [9000 + [9000 ~ 24250 1.44 MHz N] 24173] MHz N = 0:15173 FR224250 ~ 24250.08 MHz + [24174 + [24174 ~ 100000 N * 17.28 MHz, N] 28557]MHz N = 0:4383

The NR standard has two major frequency ranges (FR) specified in 3GPP.One is commonly referred to as sub-6 GHz and corresponds to frequencyrange FR1 in Tables 7 and 8 and the other is referred to as millimeterwave and corresponds to frequency range FR2 in Tables 7 and 8. Dependingon the frequency range, the maximum bandwidth and the availablesubcarrier spacing(s) differ.

Table 7 shows the channel raster, i.e., the NR-ARFCH definition, andTable 8 shows the synchronization raster.

The channel raster defines a set of RF reference frequencies that areused to identify radio frequency (RF) channel positions. The RFreference frequencies for RF channels are mapped to resource elements onthe carrier. A global frequency raster is defined for all frequenciesfrom 0 to 100 GHz and is used to define a set of allowed RF referencefrequencies. The granularity of the global frequency raster isΔF_(Global). For each operating band, a subset of frequencies from theglobal frequency raster is applicable to that band and forms a channelraster for that band with the granularity ΔF_(Global).

The synchronization raster represents the frequency position of asynchronization (SS) block that can be used by the UE for systemacquisition when there is no explicit signaling about the SS blockposition. A global synchronization raster is defined for all frequenciesand the frequency position of the SS block is defined as SSREF with thecorresponding global synchronization channel number (GSCN).

Mapping between the synchronization raster and the correspondingresource elements of the SS block is given in Table 8. The mappingdepends on the total number of RBs allocated in the channel and isapplied to both UL and DL. Table 8 shows the position of resourceelement (RE) #0 (i.e., subcarrier #0) of RB #10 of the SS block. The SSblock is composed of twenty RBs. When twenty RBs constituting the SSblock are indexed from 0 to 19, the frequency indicated by thesynchronization raster corresponds to the first RE, i.e., the positionof the first subcarrier of RB #10 among RB #0 to RB #19.

The channel raster and the SS raster are fixed to certain values asshown in Tables 7 and 8. Therefore, if the carrier frequency isexpressed as f_(Tx)=N_(Tx)*Δf+Δ_(offset), then the offset termΔ_(offset) may be limited to some specific values (e.g., −5 kHz, 0 or 5kHz) for frequency range FR1 (<3 GHz) and be 0 for the remainingfrequency bands. In the above expression, as discussed in regards toEquation 12, the term Δf is the subcarrier spacing, the term N_(Tx) is apositive integer closest to f_(Tx)/Δf (e.g., floor{f_(Tx)/Δf} orceil{f_(Tx)/Δf}), and the term Δ_(offset) is a real number whosemagnitude is smaller than Δf (hereinafter, descriptions that are appliedto f_(Tx) are also applicable to f_(Rx)).

In addition, the numbers of samples, including the number of samples fora cyclic prefix (CP), which are used for each symbol in the currentLTE/NR communication system are integer multiples of 16 for everysubcarrier spacing with respect to a sample time determined based on Δf.That is, the CP length is 144=16*9 or 160=16*10, and the length of thesignal part of the OFDM symbol other than the CP part is 2048=16*128.For example, for a bandwidth of 20 MHz, which is the 0.61=15 kHzsubcarrier spacing in the LTE or NR standard, the sampling frequency is30.72 MHz, one subframe (or one slot) is composed of 30720 samples, andeach OFDM symbol is composed of 2048+144 sample times or 2048+160 sampletimes. For reference, in the description of the present disclosure, eachsample time T_(s) is 1/(30.72 MHz), i.e., T_(s)=1/(2048*15*10³ kHz).

In some scenarios, NR and LTE systems may use values that areproportional to the numerology corresponding to the bandwidth of 20 MHz,which is a Δf=15 kHz subcarrier spacing, as numerologies, and thereforeit should be noted that all frequencies described in the presentdisclosure are based on the numerology corresponding to the bandwidth of20 MHz, which is a 15 kHz subcarrier spacing. Here, 2048 is the signallength (i.e., the effective symbol length of the OFDM symbol) defined bythe FFT size when the numerology above (e.g., the 15 kHz subcarrierspacing and the 20 MHz bandwidth) is used, 144 and 160 correspond tocyclic prefix (CP) lengths when the numerology above (e.g., the Δf=15kHz subcarrier spacing and the 20 MHz bandwidth) is used.

The phase reset for the transmitted signal and the received signal maybe implemented, for example, to address scenarios where the signalperiodicity according to the upconversion frequency is not an integermultiple of the OFDM symbol length, where the OFDM symbol length isequal to the length of the cyclic prefix (CP) part plus the length ofthe signal part. Therefore, in some scenarios, if a carrier frequencyaccording to which an integer multiple of the signal periodicity has aperiodicity corresponding to the OFDM symbol length is used, then phasereset may not be implemented.

For example, consider a communication system in which the cyclic prefix(CP) part of an OFDM symbol consists of 144=16*9 samples or 160=16*10samples, and the signal part of the OFDM symbol consists of 2048=16*128samples, and the IFFT/FFT size is 2048. In such a system, no phase resetwould be required if the upconversion frequency is set to a frequencywith a periodicity of 16 samples, or frequency 1/(16*T_(s)), where 16 isthe greatest common divisor of {144, 160, 2048}. Substituting for thesampling time T_(s), then this corresponds to an upconversion frequencyequal to 1/(16*T_(s))=1/{16*1/(FFT size*Δf)}=1/{16*1/(2048*Δf)}=128Δf.As such, if the upconversion frequency is set to the value1/(16*T_(s))=128Δf, then no phase reset would be required.

The reason for this is that a frequency having a periodicity of 16samples (16 being the greatest common divisor of 144, 160, and 2048) hasthe same phase at the beginning of each signal part of the OFDM symbols.In particular, this is because for a sine wave having a period of16*T_(s), nine such sine waves are included in a CP part having a lengthof 144T_(s), 10 such sine waves are included in a CP part having alength of 160 T_(s), and 128 such sine waves are included in a CP parthaving a length of 2048T_(s). For example, considering the minimumsubcarrier spacing Δf=15 kHz supported by the NR system, if a carrierfrequency corresponding to an integer multiple of 15 kHz*2048/16=15kHz*128=1.92 MHz is used, then the phase naturally becomes 0 at thebeginning of the signal part of each OFDM symbol, and therefore problemsof phase offset do not occur.

Further, according to a numerology having a subcarrier spacing of Δf=15kHz, cyclic prefix (CP) lengths of 144T_(s) and 160T_(s), and a signalpart (i.e., effective symbol) length 2048T_(s) of an OFDM symbol, thephase would be 0 even at the CP start point in the case of a frequencycorresponding to an integer multiple of 1.92 MHz. In more general terms,when a plurality of CP lengths (e.g., N_(CP,1), N_(CP,2), . . . ) aredefined for OFDM symbol signal generation and the number of effectivesamples per OFDM symbol, i.e., the number of samples (i.e., IFFT/FFTsize) in the signal part except the CP in an OFDM symbol, is N_(sample),then a frequency that does not cause phase discontinuity per symbolwould be a frequency of which one period corresponds to samples thenumber of which corresponds to the greatest common divisor of {N_(CP,l),N_(CP,2), . . . , N_(sample)}.

Such a frequency that does not cause phase discontinuity per symbol maybe expressed using a subcarrier spacing as follows:

${N_{base}\Delta\; f} = {\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}\Delta\;{f.}}$Here, gcd{N_(CP,1), N_(CP,2), . . . , N_(sample)} is the greatest commondivisor of N_(CP,l), N_(CP,2), . . . , and N_(sample).

Applying this to the aforementioned numerology of 2048, 160 and 144yields N_(base)Δf=128Δf. If the FFT size=4096 is used, then the cyclicprefix (CP) lengths are changed to 144*2 and 160*2 in the NR standard.Thus, applying the changed CP lengths to

${N_{base}\Delta\; f} = {\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}\Delta\; f}$yields the same result of N_(base)=128.

As another example, even when the FFT size is reduced from 2048 to, forexample, FFT size=1024, the CP lengths are changed to 144/2=72 and160/2=80. Therefore, applying the changed CP lengths to

${N_{base}\Delta\; f} = {\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}\Delta\; f}$yields the same result of N_(base)=128.

Hereinafter, the present disclosure describes implementations ofcommunication systems (e.g., LTE system, or an NR system) in which thelength of the cyclic prefix (CP) part and the length of the signal partare 144*2^(μ) or 160*2^(μ) and the length of the signal part of the OFDMsymbol is 2048*2^(μ) (where μ is an integer), as an example.

In such scenarios, a frequency that does not cause phase discontinuitymay be represented using 128Δf. However, implementations are not limitedthereto, and the present disclosure is applicable even in scenarioswhere a CP length and a signal part length different from theillustrated CP length and signal part length are used.

For example, implementations of the present disclosure may be applied inscenarios where a frequency having one period corresponding to sampleswhose number corresponds to the greatest common divisor of {N_(CP,1),N_(CP,2), . . . , N_(sample)} is used as a base carrier frequencyf_(base), that is, where an integer multiple of

$\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}\Delta\; f$is used as f_(base).

As such, in the following description, the base carrier frequencyf_(base), which was described above as an integer multiple of 128Δf, maybe generalized to an integer multiple of a frequency having one periodcorresponding to a number of samples which corresponds to the greatestcommon divisor of {N_(CP,1), N_(CP,2), . . . , N_(sample)}, or aninteger multiple of

$\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}\Delta\;{f.}$

For the scenario of using a base carrier frequency of 128Δf, the f_(Tx)can be expressed as follows:f _(Tx) =N _(Tx) ·Δf+Δ _(offset) =N _(int)·128Δf+N _(frac) ·Δf+Δoffset=f_(base) +f _(frac)+Δ_(offset)  Equation 17

Here, the term N_(int)=└N_(Tx)/128┘, the term N_(frac)=modulo(N_(Tx),128), the term f_(base) is the quantized (down-quantized, e.g., with afloor function) version obtained with a resolution of 128Δf (e.g., 1.92MHz resolution when Δf=15 kHz) among the carrier frequencies, and theterm f_(frac) is a version quantized with Δf for differences betweenf_(base) and f_(Tx).

The term Δ_(offset) represents the offset from the frequency in units ofΔf=15 kHz. In the NR system, Δ_(offset) may be set to be +/−5, 0 kHz,for example. In particular, Δ_(offset) may be defined to be one of −5kHz, 5 kHz, and 0 kHz based on the Δf=15 kHz subcarrier spacing. In someimplementations, N_(int) may be replaced by the round function insteadof the floor function. In this case, N_(frac) may be defined asN_(frac)=N_(Tx)−128*round(N_(Tx)/128). When N_(int) is replaced by theround function instead of the floor function, other operations exceptN_(frac)=N_(Tx)−128*round(N_(Tx)/128) are the same as when N_(int) isdefined as the floor function.

In Equation 17, the base carrier frequency f_(base) is a frequency foralways resetting the phase to a certain value in an OFDM symbol unit.Therefore, the expression Ψ_(Tx,l)=−2πf_(Tx)(t_(start,l)+N_(CP,l)^(μ)T_(c)) corresponding to phase compensation has the same value asΨ_(Tx,l)=−2π(N_(frac)Δf+Δ_(offset))·(t_(start,l)+N_(CP,l) ^(μ)T_(c)).

Considering only the normal cyclic prefix (CP), the value of the phasecompensation termΨ_(Tx,l)=−2π(N_(frac)Δf+Δ_(offset))·(t_(start,l)+N_(CP,l) ^(μ)T_(c)) tobe applied to one symbol for the carrier frequency difference betweenthe transmitter and the receiver is one of 128*3 complex valuesaccording to combinations of N_(frac)=0, . . . , 127 and Δ_(offset)=−5kHz, 0 kHz, 5 kHz, and is one of 128 complex values (e.g., N_(frac)=0, .. . , 127) for frequency range FR1 (>3 GHz) with Δ_(offset)=0 orfrequency range FR2. The phase compensation values to be applied to aplurality of symbols constituting a predetermined time unit (e.g., slot,subframe, 1 ms, etc.) with respect to a carrier frequency differencebetween the transmitter and the receiver differ among the symbols.

Therefore, if a set of phase compensation values for the plurality ofsymbols is expressed as a sequence, considering only the normal cyclicprefix (CP), the phase compensation termΨ_(Tx,l)=−2π(N_(frac)Δf+Δ_(offset))·(t_(start,l)+N_(CP,l) ^(μ)T_(c))needs 128*3 sequences according to the combinations of N_(frac)=0, . . ., 127 and Δ_(offset)=−5 kHz, 0 kHz, 5 kHz. If the number of possiblevalues of Δ_(offset) is changed to b other than 3, then 128*b sequencesmay be needed for phase compensation. For the frequency range FR1 (>3GHz) with Δ_(offset)=0 or FR2, only 128 sequences (e.g., N_(frac)=0, . .. , 127) may be needed. Here, the phase compensation term has a periodof up to 1 ms. That is, assuming that one period of the signal part ofeach OFDM symbol is 2048 sample times, if the phase value for a specificcarrier frequency at an arbitrary OFDM symbol boundary is α, the samephase value α appears after 2048 sample times, i.e., 1 ms, since thesample time T_(s)=1/15000*2048 seconds.f_(Tx)=N_(int)*128Δf+N_(frac)Δf+Δ_(offset), and phase compensation isnot needed for N_(int)*128Δf. Therefore, in the NR system, the phasecompensation term Ψ_(Tx,l)=−2πf_(Tx)(t_(start,l)+N_(CP,l) ^(μ)T_(c)) canbe calculated as follows.Ψ_(Tx,l)=−2π(N _(frac) Δf+Δ _(offset))·(t _(start,l) +N _(CP,l) ^(μ) T_(c))  Equation 18

Using the original definition of carrier frequencyf_(Tx)=N_(Tx)*Δf+Δ_(offset), the final signal at the transmitting sidecan be given as follows.

$\begin{matrix}{{x^{({p,\mu})}(t)} = {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot \{ {e^{j\; 2\;{\pi{({k + k_{0} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot e^{j\;\Psi_{{Tx},l}}} \} \cdot e^{j\; 2\pi\; f_{Tx}t}}}} & {{Equation}\mspace{14mu} 19}\end{matrix}$

Implementation 1

In Implementation 1, the carrier frequency f_(Tx) is used for frequencyupconversion using a free-running OSC and the carrier frequency f_(Rx)is used for frequency downconversion using the free-running OSC.

FIGS. 5A to 5C are diagrams illustrating examples of Implementation 1 ofthe present disclosure. Particularly, FIG. 5A shows an example of a partof the transmitting side structure according to Implementation 1, andFIGS. 5B and 5C show examples of a part of the receiving side structureaccording to Implementation 1.

Referring to FIG. 5A, before up-converting the OFDM baseband signal tothe carrier frequency, the transmitting side performs multiplication onthe signal at each symbol by using one complex-valued sequence (i.e.,performing phase reset) calculated with respect to the carrier frequencyf_(Tx) among the 128 complex-valued sequences or among the 128*3complex-valued sequences. Then, the transmitting side performsupconversion using f_(Tx). One of the 128 complex-valued sequences (orone of 128*3 complex-valued sequences) for the carrier frequency f_(Tx)is used for phase compensation, and a plurality of elements constitutingthe corresponding complex-valued sequence is applied to a plurality ofOFDM symbols in one-to-one correspondence.

Implementation 1 is performed in an analogous way at the receiving side.An example of an operation of Implementation 1 at the receiving side isdescribed in detail below.

In scenarios where the carrier frequency is expressed asf_(Rx)=NR_(x)*Δf+Δ_(offset), then the Δ_(offset) may be −5 kHz, 0 kHz or5 kHz in frequency range FR1 (<3 GHz) at the receiver and is 0 kHz inthe other frequency bands, as described above for the transmitterstructure. In addition, the numbers of samples, including the number ofsamples for a cyclic prefix (CP), which are used for each symbol in thecurrent LTE/NR communication system are integer multiples of 16 forevery subcarrier spacing with respect to a sample time determined basedon Δf. Therefore, in this case, when Δf=15 kHz, if a carrier frequencycorresponding to an integer multiple of 15 kHz*2048/16=15 kHz*128=1.92MHz is used for downconversion, then the phase of the signal part (i.e.,valid symbol) will naturally start with 0 at every OFDM symbol, andtherefore the aforementioned issue may not be raised. In addition, givena subcarrier spacing of Δf=15 kHz, CP lengths of 160T_(s) and 144T_(s),and an effective symbol length of 2048 T_(s), which correspond to thenumerology currently available in the LTE and NR systems, the phase is 0even at the CP start point in the case of a frequency corresponding toan integer multiple of 1.92 MHz.

More generally, when a plurality of cyclic prefix (CP) lengths (e.g.,N_(CP,l), N_(CP,2), . . . ) is defined for OFDM symbol signal generationand the number of effective samples per OFDM, i.e., the number ofsamples (i.e., IFFT/FFT size) in the signal part except the CP in anOFDM symbol, is N_(sample), then a frequency of which one periodcorresponds to samples the number of which corresponds to the greatestcommon divisor of {N_(CP,l), N_(CP,2), . . . , N_(sample)} is afrequency that does not cause phase discontinuity per symbol. Such afrequency that does not cause phase discontinuity per symbol may beexpressed using a subcarrier spacing, as follows:

${N_{base}\Delta\; f} = {\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}\Delta\;{f.}}$Here, gcd{N_(CP,1), N_(CP,2), . . . , N_(sample)} is the greatest commondivisor of N_(CP,1), N_(CP,2), . . . , and N_(sample).

Applying this to the aforementioned numerology (i.e., 2048, 160, 144)yields N_(base)Δf=128Δf. If the FFT size=4096 is used in the example ofthe bandwidth of 20 MHz, which is the Δf=−15 kHz subcarrier spacing, thecyclic prefix (CP) lengths are changed to 144*2 and 160*2 in the NRstandard. Thus, applying the changed CP lengths to

${N_{base}\Delta\; f} = {\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}\Delta\; f}$yields the same result of N_(base)=128. As another example, even whenthe FFT size is reduced from 2048 to, for example, FFT size=1024, the CPlengths are changed to 144/2=72 and 160/2=80. Therefore, applying thechanged CP lengths to

${N_{base}\Delta\; f} = {\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}\Delta\; f}$yields the same result of N_(base)=128. As mentioned above, thefrequency not causing phase discontinuity will be represented using128Δf below.

Using 128Δf, the f_(Rx) can be expressed as:f _(Rx) =N _(Rx) ·Δf+Δ _(offset) =N _(int)·128Δf+N _(frac) ·Δf+Δ_(offset) =f _(base) +f _(frac)+Δ_(offset)  Equation 20

Here, N_(int)=└N_(Rx)/128┘, N_(frac)=modulo(N_(Rx),128), f_(base) is thequantized (down-quantized, i.e., with a floor function applied) versionobtained with a resolution of 128Δf (e.g., 1.92 MHz resolution whenΔf=15 kHz) among the carrier frequencies, and f_(frac) is a versionquantized with Δf for differences between f_(base) and f_(Rx). TheΔ_(offset) represents the amount off the frequency in units of Δf=15kHz. In the NR system, Δ_(offset) may be set to be +/−5, 0 kHz, forexample. In particular, Δ_(offset) may be defined to be one of −5 kHz, 5kHz, and 0 kHz based on the Δf=15 kHz subcarrier spacing. In someimplementations, the N_(int) may be replaced by the round functioninstead of the floor function. In this case, N_(frac) can be defined asN_(frac)=N_(Tx)−128*round(N_(Tx)/128). When N_(int) is implemented bythe round function instead of the floor function, then other operationsexcept N_(frac)=N_(Tx)−128*round(N_(Tx)/128) are the same as whenN_(int) is defined as the floor function.

In Equation 20, f_(base) is a frequency for always resetting the phaseto a certain value in an OFDM symbol unit, and thereforeΨ_(Rx,l)=2πf_(Rx)(t_(start,l)+N_(CP,l) ^(μ)T_(c)) corresponding to phasecompensation has the same value asΨ_(Rx,l)=2π(N_(frac)Δf+Δ_(offset))·(t_(start,l)+N_(CP,l) ^(μ)T_(c)).Therefore, considering only the normal cyclic prefix (CP), the phasecompensation termΨ_(Rx,l)=2π(N_(frac)Δf+Δ_(offset))·(t_(start,l)+N_(CP,l) ^(μ)T_(c))needs only 128*3 sequences according to combinations of N_(frac)=0, . .. , 127 and Δ_(offset)=−5 kHz, 0, 5 kHz. If the number of possiblevalues of Δ_(offset) is changed to a value b other than 3, then 128*bsequences may be implemented for phase compensation. For frequency rangeFR1 (>3 GHz) with Δ_(offset)=0 or frequency range FR2, only 128sequences (e.g., N_(frac)=0, . . . , 127) may be implemented. Here, thephase compensation term has a period of up to 1 ms.f_(Tx)=N_(int)*128Δf+N_(frac)*Δf+Δ_(offset), and phase compensation isnot implemented for N_(int)*128Δf. Therefore, in the NR system, thephase compensation term Ψ_(Rx,l)=2πf_(Rx)(t_(start,l)+N_(CP,l)^(μ)T_(c)) may be calculated as follows.Ψ_(Rx,l)=2π(N _(frac) Δf+Δ _(offset))·(t _(start,l) +N _(CP,l) ^(μ) T_(c))  Equation 21

When the receiving side operation is defined using the originaldefinition of carrier frequency f_(Rx)=N_(Rx)*Δf+Δ_(offset), then thereceiving side operation according to Implementation 1 may be expressedas Equation 22 below.

${\overset{˜}{a}}_{k,l}^{({p,\mu})} = {e^{j\Psi_{{Rx},l}} \cdot {\int{{( {{y^{({p,\mu})}(t)}\ .\ e^{{- j}2\pi f_{Rx}t}} ) \cdot e^{{- j}\; 2\;{\pi{({k + k_{0,{Rx}} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}}}d\; t}}}$

In Equation 22, the integral represents a function corresponding to theFFT, and the operation of Equation 22 may be expressed as shown in FIG.5B. When analog-to-digital conversion is performed after the receivedsignal is actually down-converted, the FFT expressed by the integralequation is implemented in the form of a discrete equation like Equation23. The operation of Equation 23 may be expressed as shown in FIG. 5C.

$\begin{matrix}{{\overset{\sim}{a}}_{k,l}^{({p,\mu})} = {e^{j\;\Psi_{{Rx},l}} \cdot {\sum\limits_{i}\{ {( {{y^{({p,\mu})}(t)} \cdot e^{{- j}\; 2\pi\; f_{Rx}t} \cdot {\delta( {t - {i \cdot T_{c}}} )}} ) \cdot e^{{- j}\; 2{\pi{({k + k_{0,{Rx}} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}\Delta\;{f{({{i \cdot T_{c}} - {({t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}})}}}} \}}}} & {{Equation}\mspace{14mu} 23}\end{matrix}$

The difference between the examples of FIG. 5B and FIG. 5C is theposition of the phase reset function, with other functions beingequivalent.

After performing downconversion on the received signal using f_(Rx) orperforming downconversion and FFT on the received signal using f_(Rx),the receiving side performs phase reset of performing multiplication onthe signal at every symbol by using one complex-valued sequencecalculated by the carrier frequency f_(Rx) among the 128 complex-valuedsequences or among the 128*3 complex-valued sequences.

The transmitting device and the receiving device according toImplementation 1 may store

$\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}$or

$\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}*( {{number}\mspace{14mu}{of}\mspace{14mu}\Delta_{offset}} )$sequences, or

$\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}$or

$\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}*( {{number}\mspace{14mu}{of}\mspace{14mu}\Delta_{offset}} )$sequences (corresponding to the value of 2π(N_(frac)Δf+Δ_(offset)) fromΨ_(Rx,l)=2π(N_(frac)Δf+Δ_(offset))·(t_(start,l)+N_(CP,l) ^(μ)T_(c))excluding (t_(start,l)+N_(CP,l) ^(μ)T_(c))), and use the same uponperforming phase reset for each symbol. (t_(start,l)+N_(CP,l) ^(μ)T_(c))has certain values for a subcarrier spacing. Therefore, if the values of2π(N_(frac)Δf+Δ_(offset)) are fixed to the

$\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}$or

$\frac{N_{sample}}{gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}*( {{number}\mspace{14mu}{of}\mspace{14mu}\Delta_{offset}} )$sequences according to Implementation 1 of the present disclosure, thephase compensation for the corresponding symbol on a symbol-by-symbolbasis may be performed by simply selecting one of the sequences, andtherefore phase reset may be easily implemented in the transmittingdevice and the receiving device.

As such, according to some implementations of the present disclosure,when the carrier frequency is changed, the sequence for phasecompensation may also be changed, but the changed sequence is one ofonly 128 or 128*3 possible sequences. Therefore, in some implementationsof the present disclosure, each of the transmitting side and thereceiving side may store a sequence composed of phase compensationvalues to be applied to OFDM symbols corresponding to a positive integermultiple of a period in which phase changes, for each available carrierfrequency, and perform phase compensation by applying a sequencecorresponding to a specific carrier frequency in every period duringprocessing of OFDM symbol signals using the specific carrier frequency.

For example, if the phase of the OFDM symbols changes with a periodicityof 1 ms and 14 OFDM symbols are included in the 1 ms period, then thephase compensation value sequence for a specific carrier frequency iscomposed of 14 phase compensation values for each of the 14 OFDMsymbols. The transmitting side and the receiving side may store a phasecompensation value sequence to be applied at intervals of 1 ms for eachcarrier frequency, and may use a stored phase compensation valuesequence to perform phase compensation on the corresponding carrierfrequency.

Implementation 2.

Similar to Implementation 1, Implementation 2 also uses a base carrierfrequency of f_(base)=N_(int)*128Δf to facilitate performing phase resetor to facilitate addressing phase mismatch between OFDM symbols. Here,the term f_(base) is a frequency that is closest to f_(XX) (where thesubscript XX is Tx for the transmitting and Rx for the receiving side)among the frequencies which are integer multiples of 128Δf (e.g., amongfrequencies less than or equal to f_(XX) or among frequencies greaterthan or equal to f_(XX) or among frequencies on both sides of f_(TX)).Hereinafter, f_(base) is represented by N_(int)*128Δf (where N_(int) isan integer).

By way of comparison, in Implementation 1 described above, the carrierfrequency f_(Tx) was used for frequency upconversion using afree-running OSC (i.e., an analog OSC) and the carrier frequency f_(Rx)is used for frequency downconversion using the free-running OSC.

By contrast, in Implementation 2, the base carrier frequencyf_(base)=N_(int)*128Δf of f_(XX) is used to perform frequency shift(e.g., frequency shift with a free-running OSC) at the analog stage, andthe frequency difference ‘f_(XX)−N_(int)*128Δf’ of f_(XX) is used toperform frequency shift at the digital stage. When N_(int)*128Δf is usedas the upconversion/downconversion frequency in the free-running OSC,the same phase shift value is given for the OFDM symbols (i.e., thephases at the beginning of the signal parts of the OFDM symbols are thesame). Therefore, when N_(int)*128Δf is used as theupconversion/downconversion frequency in the free-running OSC, it maynot be necessary to calculate and apply a phase shift value per OFDMsymbol for phase compensation.

Hereinafter, two examples of Implementation 2 will be described asImplementation 2-1 and Implementation 2-2 according to a module to beused to frequency-shift ‘f_(XX)−N_(int)*128Δf’ in the digital stage.

Implementation 2-1

FIGS. 6A and 6B are diagrams illustrating examples of Implementation 2-1of the present disclosure. Particularly, FIG. 6A shows an example of apart of the transmitting side structure according to Implementation 2-1,and FIG. 6B shows an example of a part of the receiving side structureaccording to Implementation 2-1. In FIGS. 6A and 6B, the t_(l) denotesthe start position of the signal part of an OFDM symbol l in the timedomain, and may be expressed as t_(l)=t_(start,l)+N_(CP,l) ^(μ)T_(c).

When the final transmitter signal is summarized using Equation 17, thetransmitted signal of the transmitter can be expressed by the followingequation.

$\begin{matrix}\begin{matrix}{{x^{({p,\mu})}(t)} = {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ e^{{{j2\pi}{({k + k_{0} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \} \cdot} \\{e^{{j2\pi f}_{Tx}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ {e^{{{j2\pi}{({k + k_{0} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} } \\{e^{{j2\pi fN}_{frac}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} \\{ e^{{{j2\pi}\Delta}_{offset}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}} \} \cdot} \\{e^{{j2\pi f}_{base}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ {e^{{{j2\pi}{({k + k_{0} + N_{frac} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} } \\{ e^{{{j2\pi}\Delta}_{offset}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}} \} \cdot e^{{j2\pi f}_{{base}^{t}}}}\end{matrix} & {{Equation}\mspace{20mu} 24}\end{matrix}$

Referring to the last line of Equation 24, N_(frac)=modulo(N_(Tx),128)is a term for changing a mapping position of a resource (i.e., a mappingposition of a subcarrier) in an IFFT term. That is, in the term

e^(j 2 π (k + k₀ + N_(frac) − N_(grid)^(size, μ)N_(sc)^(RB)/2)Δ f(t − t_(start, l) − N_(CP, l)^(μ)T_(c)))corresponding to the IFFT term, the term k+k₀+N_(frac)−N_(grid)^(size,μ)N_(sc) ^(RB)/2 denotes the mapping position of a resource forIFFT. Thus, the frequency to which the signal symbol a_(k) of eachsubcarrier is to be modulated is determined depending on the value ofk+k₀+N_(frac)−N_(grid) ^(size,μ)N_(sc) ^(RB)/2.

As such, in Implementation 2-1, the baseband signal is frequency-shiftedby ‘f_(Tx)−N_(int)*128Δf’ (or a part of ‘f_(Tx)−N_(int)*128Δf’ ifΔ_(offset) is not 0) of f_(Tx) by changing the mapping position of theresource with respect to the IFFT by N_(frac). Since the IFFT itself hasa function to reset the phase, the frequency shift performed by changingthe resource mapping position with respect to the IFFT does not causephase mismatch between the OFDM symbols. The term

e^(j 2 πΔ_(offset)(t − t_(start, l) − N_(CP, l)^(μ)T_(c)))in the last line of Equation 24 resets the phase of the signal to acertain value (e.g., 0) at the start or end of the cyclic prefix (CP) ona symbol-by-symbol basis and frequency-shifts the signal by Δ_(offset),which is similar to 7.5 kHz frequency shift (see ½*Δf in Equation 1)performed in LTE uplink SC-FDMA.

In some scenarios, it is generally difficult to digitally implement avery large frequency shift. By contrast, frequency shift by Δ_(offset)may be easily implemented by a digital OSC since the value of Δ_(offset)is small. The frequency shift by Δ_(offset) is performed after IFFT. Inthe last line of Equation 24, the frequency f_(base) in the expressione^(j2πf) ^(base) ^(t) is a frequency that is closest to f_(TX) among thefrequencies corresponding to integer multiples of 128Δf (e.g., amongfrequencies less than or equal to f_(Tx) or among frequencies greaterthan or equal to f_(Tx) or among frequencies on both sides of f_(TX)).As examples, the frequency f_(base) may be integer multiples of 1.92 MHzwhen Δf=15 kHz and may be integer multiples of 3.84 MHz when Δf=30 kHz.The expression e^(j2πf) ^(base) ^(t) represents the operation ofupconversion to f_(base). In Implementation 2-1, the frequency shift byf_(base) may be performed using an analog OSC.

Implementation 1, described above, configured the phase compensationfunction to perform phase compensation using one of a specific number ofcomplex-valued sequences (e.g., 128 or 128*3 complex-valued sequences).In particular, in Implementation 1, described above, frequencyupconversion to the carrier frequency f_(Tx), was performed by an analogOSC. By contrast, in Implementation 2-1, frequency up-shift to thecarrier frequency f_(Tx) is actually performed through frequency shiftof (f_(Tx)−N_(int)*128Δf) using IFFT, which is a digital module, andfrequency shift of N_(in)*128Δf using an analog OSC.

Referring to FIG. 6A, considering Δ_(offset), a frequency shiftoperation corresponding to Δ_(offset) may be performed in addition tothe IFFT and upconversion functions. In some implementations, Δ_(offset)may have a variance of +/−5 kHz, although implementations are notlimited thereto. In addition, in the case of frequency ranges FR2 andFR1 (>3 GHz) or in the case of Δ_(offset)=0 in frequency range FR1 (<3GHz), if Δ_(offset)=0, the frequency shift operation may be performedwithout any additional operation (e.g., frequency shift by Δ_(offset)).In this case, the frequency shift module denoted by e^(j2πΔ) ^(offset)^((t−t) ^(l) ⁾ in FIG. 6A may be omitted.

Corresponding operations may be configured at the receiving side. Thereceiving side operation according to Implementation 2 will be describedin detail below.

The final receiver signal summarized using Equation 20 may be given asfollows.

$\begin{matrix}\begin{matrix}{{\overset{\sim}{a}}_{k,l}^{({p,\mu})} = {\sum\limits_{i}\{ {( {{{y^{({p,\mu})}(t)} \cdot e^{- {{j2\pi f}_{Rx}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot \delta}( {t - {i \cdot T_{c}}} )} ) \cdot} }} \\ e^{{- {{j2\pi}{({k + k_{0,{Rx}} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}}{{\Delta f}{({{i \cdot T_{c}} - {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}})}}} \} \\{= {\sum\limits_{i}\{ {( {{{y^{({p,\mu})}(t)} \cdot e^{- {{j2\pi f}_{base}{({t - l_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot \delta}( {t - {i \cdot T_{c}}} )} ) \cdot} }} \\{e^{{- {{j2\pi}{({k + k_{0,{Rx}} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}}{{\Delta f}{({{i \cdot T_{c}} - {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}})}}} \cdot} \\{ e^{{- {j2\pi N}_{frac}}{{\Delta f}{({{i \cdot T_{c}} - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \} \cdot} \\{e^{- {{{j2\pi}\Delta}_{offset}{({{i \cdot T_{c}} - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}}} \\{= {\sum\limits_{i}\{ {( {{y^{({p,\mu})}(t)} \cdot e^{- {j2\pi f}_{{base}^{t}}} \cdot {\quad{\delta( {t - {i \cdot T_{c}}} )}}} ) \cdot} }} \\{ e^{{- {{j2\pi}{({k + k_{0,{Rx}} + N_{frac} - N_{grid}^{{size},\mu} - {N_{sc}^{RB}/2}})}}}{{\Delta f}{({{i \cdot T_{c}} - {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}})}}} \} \cdot} \\{e^{- {{{j2\pi}\Delta}_{offset}{({{i \cdot T_{c}} - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}}}\end{matrix} & {{Equation}\mspace{14mu} 25}\end{matrix}$

Referring to the last line of Equation 25, N_(frac)=modulo (N_(Rx),128)is a term for changing the de-mapping position of a resource in the FFTterm. That is, in the FFT expression

e^(−j 2 π (k + k_(0, Rx) + N_(frac) − N_(grid)^(size, μ)N_(sc)^(RB)/2)Δ f(t − t_(start, l) − N_(CP, l)^(μ)T_(c))),the term k+k_(0,Rx)+N_(frac)−N_(grid) ^(size,μ)N_(sc) ^(RB)/2 is a termrelating to the de-mapping position of a resource, i.e., the de-mappingposition of a subcarrier to an OFDM baseband signal. In other words,k+k_(0,Rx)+N_(frac)−N_(grid) ^(size,μ)N_(sc) ^(RB)/2 is a term relatingto the output positions of the subcarriers from the FFT. InImplementation 2-1, frequency shift is performed by‘f_(Rx)−N_(int)*128Δf’ (or ‘f_(Tx)−N_(int)*128Δf’ if Δ_(offset) is not0) of f_(Rx) by changing the de-mapping position of the resource withrespect to the FFT through N_(frac). Since the FFT itself has a functionto reset the phase, the frequency shift performed by changing theresource de-mapping position with respect to the FFT does not causephase mismatch between the OFDM symbols. The expression

e^(−j 2 πΔ_(offset)(i ⋅ T_(c) − t_(start, l) − N_(CP, l)^(μ)T_(c)))in the last line of Equation 25 resets the phase of the signal to acertain value (e.g., 0) at the start or end of the cyclic prefix (CP) ona symbol-by-symbol basis and frequency-shifts the signal by Δ_(offset),which is similar to 7.5 kHz frequency shift (see ½*Δf in Equation 1)performed in LTE uplink SC-FDMA.

In some scenarios, it may be difficult to digitally implement a verylarge frequency shift. By contrast, frequency shift by Δ_(offset) may beeasily implemented by a digital OSC since the value of Δ_(offset) issmall. The frequency shift by Δ_(offset) is performed before FFT. In thelast line of Equation 25, the frequency f_(base) in the expressione^(−j2πf) ^(base) ^(t) is a frequency that is closest to f_(Rx) amongthe frequencies corresponding to integer multiples of 128Δf (e.g., amongfrequencies less than or equal to f_(Rx) or among frequencies greaterthan or equal to f_(Rx) or among frequencies on both sides of f_(RX)).As examples, the frequency f_(base) may be integer multiples of 1.92 MHzwhen Δf=15 kHz and may be integer multiples of 3.84 MHz when Δf=30 kHz.The expression e^(−j2πf) ^(base) ^(t) represents the operation ofdownconversion from f_(base). In Implementation 2-1, the frequency shiftby f_(base) may be performed using an analog OSC.

Implementation 1, described above, configured the phase compensationfunction to perform phase compensation using one of a specific number ofcomplex-valued sequences (e.g., 128 or 128*3 complex-valued sequences).In particular, in Implementation 1, described above, frequencydownconversion from the carrier frequency f_(Rx) was performed by ananalog OSC.

By contrast, in Implementation 2-1 described herein, frequencydown-shift from the carrier frequency f_(Rx) is actually performed by afrequency shift of ‘f_(Rx)−N_(int)*128Δf’ using FFT, which is a digitalmodule, and frequency shift of f_(base)=N_(int)*128Δf using an analogOSC.

Referring to FIG. 6B, a frequency shift operation corresponding toΔ_(offset) may be performed in addition to the FFT and downconversionfunctions. In addition, in the case of frequency ranges FR2 and FR1 (>3GHz) or in the case of Δ_(offset)=0 in frequency range FR1 (<3 GHz), ifΔ_(offset)=0, then the frequency shift operation may be performedwithout any additional operation (e.g., frequency shift by Δ_(offset)).In this case, the frequency shift module denoted by e^(−j2πΔ) ^(offset)^((t-t) ^(l) ⁾ in FIG. 6B may be omitted.

Implementation 2-2

In Implementation 2-2, as in Implementation 2-1 described above,frequency shift of f_(base) of the carrier frequency f_(XX) is performedusing an analog OSC. However, in Implementation 2-2, frequency shiftingof ‘f_(XX)−f_(base)’ of f_(XX) (which was performed using IFFT/FFT inImplementation 2-1) is performed by the digital OSC.

For example, frequency shift by the digital OSC may be performed by thedigital OSC multiplying a signal by a cosine value or a sine value. Inthis case, the digital oscillator may obtain the cosine/sine value byany appropriate technique, for example by reading the cosine/sine valuefrom a computer memory, or by calculating the cosine/sine value. Inorder to make the phase of the signal have a certain value at a specificpoint in time, a digital OSC only needs to be configured such that anaddress of a memory read by the digital OSC for a specific sample is theaddress of the memory that stores a cosine/sine value that makes thephase be the certain value. Alternatively, if the digital OSC isimplemented to calculate a cosine/sine value rather than reading it fromthe memory, then the digital oscillator only needs to adjust the phaseto a desired value at a specific point in time. That is, the digital OSCmay be implemented to read a memory address storing a frequency shiftvalue having the certain phase value for a specific point in time/sampleor to adjust the phase to the certain phase value for the specific pointin time/sample. Therefore, for the frequency shift by the digital OSC,such implementations may simplify the phase reset function based on theOFDM symbol boundary. In such scenarios, some implementations may notneed to perform phase pre-compensation at the transmitting end.

For reference, phase mismatch between OFDM symbols may result from thedifference between the time interval during which IFFT/FFT is appliedand the time interval during which the free-running OSC is running.Implementing the upconversion/downconversion frequency with a digitalOSC instead of a free-running OSC, which is an analog OSC, may enableeasier and simpler resetting of the phase at the boundary of OFDMsymbols. However, upconversion/downconversion to the extent of thecarrier frequency by the digital OSC may cause a very high complexitybecause the transmitter and the receiver may need to performmultiplication in units of several GHz. For example, to performupconversion/downconversion to/from 2 GHz using a digital OSC, samplingshould be performed in units of at least 4 GHz according to the Nyquistsampling theorem. Accordingly, the digital OSC must be implemented tomultiply the input signal by the cosine/sine value ofupconversion/downconversion on the basis of the sampling unit of 4 GHz.Implementing such a large number of multiplication operations with adigital module may be very complex in some scenarios, and may lead to anincrease in the manufacturing cost of the transmitter and the receiver.Thus, in some scenarios, the overall magnitude of the carrier frequencyis not up-converted/down-converted by the digital OSC.

By way of comparison, in Implementation 2-1, described above, thetransmitting side is configured to perform a process of upconversion toa frequency corresponding to f_(base) by the OSC of the RF stage (i.e.,the free-running OSC) and a process of determining the position ofresource mapping in the IFFT using f_(frac) (or ‘f_(TX)−f_(base)’). Inscenarios where Δ_(offset) is not 0, the transmitting side according toImplementation 2-1, described above, may further perform the step ofresetting the phase on the OFDM symbol-by-symbol basis (e.g. to have thezero phase at the end time of the cyclic prefix CP) using a digital OSCfor Δ_(offset). In some scenarios of Implementation 2-1, describedabove, the signal output in the baseband may become asymmetric withrespect to DC depending on the value of f_(frac) (or ‘f_(Tx)−f_(base)’).This may limit the efficiency of the spectrum by filtering after theIFFT output in the transmitter (or before the input of the FFT in thereceiver). For example, a portion of the output of the IFFT (or an inputof FFT) may be outside of the filtering region of the transmitter (orreceiver), resulting in signal distortion at the band edges due tofiltering.

Thus, to address such problems, in some scenarios it may be desirable tomodify the transmitting side operations of Implementation 2-1 forimplementing the function of frequency upconversion (i.e., frequencyup-shift) and phase reset corresponding to f_(frac) (or‘f_(Tx)−f_(base)’) through the position of resource mapping of the IFFT.

Accordingly, to this end, in Implementation 2-2 described herein, at thetransmitting side, the function of frequency shift and phase reset for afrequency corresponding to f_(frac) (or ‘f_(Tx)−f_(base)’) areconfigured with a digital OSC as in the case of frequency shift in FIG.6A, i.e., frequency shift by Δ_(offset) using a digital OSC.

Further examples of the above-described scenarios are provided withreference to FIGS. 7A and 7B, below.

FIGS. 7A and 7B are diagrams illustrating examples of resource mappingaccording to Implementation 2-1 and Implementation 2-2 of the presentdisclosure, respectively. Specifically, FIG. 7A illustrates an exampleof resource mapping and upconversion according to Implementation 2-1,and FIG. 7B illustrates an example of resource mapping and upconversionaccording to Implementation 2-2.

Referring to the left part of FIG. 7A, in some wireless communicationsystems, information symbols a_(k) of subcarriers in an OFDM symbol(where k is a subcarrier index) are mapped to an IFFT module, and theinformation symbols mapped to the IFFT module are (approximately)symmetrically distributed with respect to the center of the IFFT moduleor the DC.

By contrast, referring to the right part of FIG. 7A, in Implementation2-1 of the present disclosure, the mapping position to the IFFT of a_(k)is shifted, for example, by an amount N_(frac). In such scenarios wherethe resource mapping position of the IFFT is changed as shown in theright part of FIG. 7A, then a portion of the output of the IFFT may beoutside of the filtering region of the transmitter. Since the portion ofthe output of the IFFT that is outside the filtering region would not besubjected to filtering, the signal may be distorted at the band edge.

In order to address this issue, as shown in FIG. 7B, Implementation 2-2shifts the frequency position of the output of the IFFT byf_(frac)+Δ_(offset) (i.e., ‘f_(Tx)−f_(base)’) via a digital oscillationto a desired position (after performing suitable filtering if any) withthe IFFT resource mapping position unchanged.

FIGS. 8A and 8B are diagrams illustrating examples of Implementation 2-2of the present disclosure. Specifically, FIG. 8A shows an example of apart of the transmitting side structure according to Implementation 2-2,and FIG. 8B shows an example of a part of the receiving side structureaccording to Implementation 2-2. In FIGS. 8A and 8B, the term t_(l)denotes the start position of the signal part of an OFDM symbol l in thetime domain, and may be expressed as t_(l)=t_(start,l)+N_(CP,l)^(μ)T_(c).

Referring first to FIG. 8A, in order to perform frequency upconversionby f_(Tx), the transmitter according to Implementation 2-2 performsupconversion for a frequency corresponding to f_(base) of f_(TX) by anoscillator (i.e., an analog OSC) in the RF stage, and performs frequencyupconversion and phase reset for a frequency corresponding tof_(frac)+Δ_(offset) (i.e., ‘f_(Tx)−f_(base)’) (e.g. so as to have thezero phase at the end time of the cyclic prefix CP) by a digital OSC onan OFDM symbol-by-symbol basis. The receiving side operations accordingto Implementation 2-2 may be performed in an analogous manner as thetransmitting side, described next.

Referring to FIG. 8B, in order to perform frequency downconversion byf_(Rx), the receiver according to Implementation 2-2 performsdownconversion for a frequency corresponding to f_(base) of f_(Rx) by anoscillator (i.e., an analog OSC) in the RF stage, and performs frequencydownconversion and phase reset for a frequency corresponding tof_(frac)+Δ_(offset) (i.e., ‘f_(Rx)−f_(base)’) (e.g. so as to have thezero phase at the end time of the cyclic prefix CP) by a digital OSC onthe OFDM symbol-by-symbol basis.

By comparison, in Implementation 2-1 described above, the receiving sideis configured to perform a process of downconversion with a frequencycorresponding to f_(base) by the OSC of the RF stage (i.e., thefree-running OSC) and a process of determining the position of resourcede-mapping in the FFT using f_(frac) (or ‘f_(Rx)−f_(base)’). If thevalue of_(offset) is not 0, then the receiving side according toImplementation 2-1, described above, may perform the process ofresetting phase on the OFDM symbol-by-symbol basis (e.g. to have thezero phase at the end time of the cyclic prefix CP) using a digital OSCfor Δ_(offset). In some scenarios of Implementation 2-1 described above,the degree of asymmetry of the signal output in the RF downconversionstep may be large with respect to DC depending on the value off f_(frac)(or ‘f_(Rx)−f_(base)’). This may limit the efficiency of the spectrum byfiltering at the downconversion output stage of the receiver.

Thus, to address such problems, in some scenarios it may be desirable tomodify the receiving side operations of Implementation 2-1 forimplementing frequency downconversion (i.e., frequency down-shift) andphase reset corresponding to f_(frac) (or ‘f_(Rx)−f_(base)’) through theposition of the resource mapping of the FFT.

Accordingly, to this end, in Implementation 2-2, at the receiving side,the function of frequency shift and phase reset for a frequencycorresponding to f_(frac) (or ‘f_(Rx)−f_(base)’) are configured with adigital OSC as in the case of frequency shift in FIG. 6B, i.e.,frequency shift by Δ_(offset) using a digital OSC. Here, the number offrequencies f_(frac) (or ‘f_(Rx)−f_(base)’) used for the frequencydownconversion and the phase reset may be 128 (or 128*3 if Δ_(offset) isnot 0 according to the frequency band).

Implementation 3

FIGS. 9A to 9C are diagrams illustrating examples of Implementation 3 ofthe present disclosure. Specifically, FIG. 9A shows an example of a partof the transmitting side structure according to Implementation 3, andFIGS. 9B and 9C show examples of a part of the receiving side structureaccording to Implementation 3.

Equation 24, above, may be modified as follows.

$\begin{matrix}\begin{matrix}{{x^{({p,\mu})}(t)} = {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ e^{{{j2\pi}{({k + k_{0} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \} \cdot} \\{e^{{{{j2\pi f}_{{Tx}(}t} - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ {e^{{{j2\pi}{({k + k_{0} + {N_{{frac} -}N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} } \\{ e^{{j2\pi\Delta}_{offset}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}} \} \cdot e^{{j2\pi f}_{base}t}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ {e^{{{j2\pi}{({k + k_{0} + N_{frac} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} } \\{ e^{- {{{j2\pi}\Delta}_{offset}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \} \cdot e^{{{j2\pi}{({f_{base} + \Delta_{offset}})}}t}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ {e^{{{j2\pi}{({k + k_{0,{Rx}} + N_{frac} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} } \\{ e^{{j\Psi}_{{Tx\_ offset},l}} \} \cdot e^{{{j2\pi}{({f_{base} + \Delta_{offset}})}}t}}\end{matrix} & {{Equation}\mspace{14mu} 26}\end{matrix}$

Based on the last line of Equation 26, Implementation 3 of the presentdisclosure will be described. Implementation 3 and Implementation 2-1have similarities in that, for example, N_(frac)=modulo(N_(Tx),128)plays the same role in Implementation 3 as in Implementation 2-1.Referring to FIG. 9A, the term N_(frac)=modulo(N_(Tx),128) is a term forchanging the mapping position of a resource (i.e., the mapping positionof a subcarrier) in the IFFT term. As described in Implementation 2-1,above,

e^(j 2 π (k + k₀ + N_(frac) − N_(grid)^(size, μ)N_(sc)^(RB)/2)Δ f(t − t_(start, l) − N_(CP, l)^(μ)T_(c)))corresponds to the IFFT term. In the IFFT term, k+k₀+N_(frac)−N_(grid)^(size,μ)N_(sc) ^(RB)/2 denotes the mapping position of a resource forIFFT.

Referring to e^(j2π(f) ^(base) ^(+Δ) ^(offset) ^()t) in the last line ofEquation 26, Implementation 3 differs from Implementation 2-1 asfollows. In the expression e^(j2π(f) ^(base) ^(+Δ) ^(offset) ^()t), thefrequency f_(base) is a frequency that is closest to f_(Tx) among thefrequencies corresponding to integer multiples of 128Δf (e.g., amongfrequencies less than or equal to f_(TX) or among frequencies greaterthan or equal to f_(Tx), or among frequencies on both sides of f_(TX)).As examples, the frequency f_(base) may be equal to an integer multipleof 1.92 MHz when Δf=15 kHz, and may be equal to an integer multiple of3.84 MHz when Δf=30 kHz, and may be equal to an integer multiple of 7.68MHz when Δf=60 kHz, and may be equal to an integer multiple of 15.36 MHzwhen Δf=120 kHz. The frequency f_(base) may be expressed asN_(int)*128Δf (where N_(int) is an integer).

The expression e^(j2π(f) ^(base) ^(+Δ) ^(offset) ^()t) represents anoperation of upconversion to a frequency obtained by adding Δ_(offset)to f_(base) (i.e., f_(base)+Δ_(offset)), and is processed by an analogOSC. Since the Δ_(offset) processed by the analog OSC may cause phasediscontinuity per symbol, in some implementations, the transmitting sideof Implementation 3 performs phase compensation for Δ_(offset) usingΨ_(Tx_offset,l)=−2πΔ_(offset)·(t_(start,l)+N_(CP,l) ^(μ)T_(c)).

By way of comparison, Implementation 2, described above, shifted thefrequency on a sample-by-sample basis by rotating the phase (i.e.,shifting the phase) on a sample-by-sample basis for frequency shift byΔ_(offset). That is, Implementation 2, described above, utilized acomplex value derived on a sample-by-sample basis to rotate the phase ona sample-by-sample basis. By contrast, Implementation 3 only multipliesthe output of the IFFT by a fixed complex value on an OFDMsymbol-by-symbol basis, as shown in FIG. 9A.

Furthermore, by way of comparison, Implementation 1, described above,utilized 128 or 128*3 complex-valued sequences for phase compensationfor all available carrier frequency candidates. That is, sinceImplementation 1 utilizes one of 128 or 128*3 complex values per symbolfor each carrier frequency, Implementation 1 utilizes 128 or 128*3complex-valued sequences for phase compensation according to carrierfrequencies. By contrast, Implementation 3 does not utilize 128 or 128*3complex-valued sequences for f_(frac) because Implementation 3 performsfrequency shift by f_(frac) by changing the resource mapping position toIFFT, and only performs phase compensation for Δ_(offset) that isimplemented through analog frequency upconversion. Therefore, sinceImplementation 3 only performs phase compensation for Δ_(offset)=−5 kHzor Δ_(offset)=5 kHz, it utilizes only two complex-valued sequences forphase compensation for Δ_(offset)=+/−5 kHz. Since the negative frequencycorresponds to the opposite phase of a positive frequency,Implementation 3 actually utilizes only one complex-valued sequence forphase compensation. In some implementations, in the case of frequencyranges FR2 and FR1 (>3 GHz) or in the case of Δ_(offset)=0 in frequencyrange FR1 (<3 GHz), the function of upconversion to the carrierfrequency may be configured without any additional operation (e.g.,frequency shift by Δ_(offset)) other than IFFT and upconversion, as inthe case of Implementation 2.

Implementation 3 is applicable to the receiving side as well as thetransmitting side. Equation 25 may be modified into the followingequation.

$\begin{matrix}\begin{matrix}{{\overset{\sim}{a}}_{k,l}^{({p,\mu})} = {\sum\limits_{i}\{ {( {{y^{({p,\mu})}(t)} \cdot e^{- {{j2\pi f}_{Rx}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot {\quad{\delta( {t - {i \cdot T_{c}}} )}}} ) \cdot} }} \\ e^{{- {{j2\pi}{({k + k_{0,{Rx}} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}}{{\Delta f}{({{i \cdot T_{c}} - {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}})}}} \} \\{= {\sum\limits_{i}\{ {( {{y^{({p,\mu})}(t)} \cdot e^{{- {j2\pi f}_{base}}t} \cdot {\quad{\delta( {t - {i \cdot T_{c}}} )}}} ) \cdot} }} \\{e^{{{- {{j2\pi}{({k + k_{0,{Rx}} + N_{frac} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}}{{\Delta f}{({{i \cdot T_{c}} - {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}})}}}\}} \cdot} \\{⁠e^{- {{{j2\pi}\Delta}_{offset}{({{i \cdot T_{c}} - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}}} \\{= {\sum\limits_{i}\{ {( {{y^{({p,\mu})}(t)} \cdot e^{- {{j2\pi}{({f_{base} + \Delta_{offset}})}}^{t}} \cdot {\quad{\delta( {t - {i \cdot T_{c}}} )}}} ) \cdot} }} \\{e^{{{- {{j2\pi}{({k + k_{0,{Rx}} + N_{frac} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}}{{\Delta f}{({{i \cdot T_{c}} - {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}})}}}\}} \cdot} \\{\cdot e^{{{j2\pi}\Delta}_{offset}{({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}}} \\{= {\sum\limits_{i}\{ {( {{y^{({p,\mu})}(t)} \cdot e^{- {{j2\pi}{({f_{base} + \Delta_{offset}})}}^{t}} \cdot {\quad{\delta( {t - {i \cdot T_{c}}} )}}} ) \cdot} }} \\{e^{{{- {{j2\pi}{({k + k_{0,{Rx}} + N_{frac} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}}{{\Delta f}{({{i \cdot T_{c}} - {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}})}}}\}} \cdot} \\{e^{- {j\Psi}_{{Rx\_ offset},l}}}\end{matrix} & {{Equation}\mspace{14mu} 27}\end{matrix}$

Based on the last line of Equation 27, Implementation 3 of the presentdisclosure will be described. Implementation 3 and Implementation 2-1have some similarities in that, for example, N_(frac)=modulo(N_(Rx),128)plays the same role in both Implementation 3 and Implementation 2-1.Referring to FIGS. 9B and 9C, the term N_(frac)=modulo(N_(Rx),128) is aterm for changing the de-mapping position of a resource (i.e., thede-mapping position to a subcarrier) in the FFT term. As previouslydescribed in regards to Implementation 2-1, above,

e^(−j 2 π (k + k_(0, Rx) + N_(frac) − N_(grid)^(size, μ)N_(sc)^(RB)/2)Δ f(t − t_(start, l) − N_(CP, l)^(μ)T_(c)))corresponds to the FFT term. In the FFT term,k+k_(0,Rx)+N_(frac)−N_(grid) ^(size,μ)N_(sc) ^(RB)/2 denotes thede-mapping position of a resource (e.g., a subcarrier) by FFT. As such,k+k_(0,Rx)+N_(frac)−N_(grid) ^(size,μ)N_(sc) ^(RB)/2 is a term relatingto the output positions of subcarriers from the FFT.

Referring to e^(−j2π(f) ^(base) ^(+Δ) ^(offset) ^()t) in the last lineof Equation 27, Implementation 3 differs from Implementation 2-1 asfollows. In the expression e^(−j2π(f) ^(base) ^(+Δ) ^(offset) ^()t), thefrequency f_(base) is a frequency that is closest to f_(Rx) among thefrequencies corresponding to integer multiples of 128Δf (e.g., amongfrequencies less than or equal to f_(Rx) or among frequencies greaterthan or equal to f_(Rx) or among frequencies on both sides of f_(Rx)).The frequency f_(base) may be expressed as N_(int)*128Δf (where N_(int)is an integer).

The expression e^(−j2π(f) ^(base) ^(+Δ) ^(offset) ^()t) represents anoperation of downconversion by the frequency obtained by addingΔ_(offset) to f_(base), and is processed by an analog OSC. SinceΔ_(offset) processed by the analog OSC causes phase discontinuity persymbol, the receiving side of Implementation 3 performs phasecompensation using Ψ_(Rx_offset,l)=2πΔ_(offset)·(t_(start,l)+N_(CP,l)^(μ)T_(c)). Similar to the description regarding FIGS. 5B and 5C, above,phase compensation usingΨ_(Rx_offset,l)=2πΔ_(offset)·(t_(start,l)+N_(CP,l) ^(μ)T_(c)) may beperformed before FFT as shown in FIG. 9B or may be performed after FFTas shown in FIG. 9C. In these examples, FIG. 9B and FIG. 9C differ onlyin the position at which phase compensation for Δ_(offset) performed,and other receiver operations/functions are the same.

By way of comparison, Implementation 2 shifts the frequency on asample-by-sample basis by rotating the phase on a sample-by-sample basisfor frequency shift by Δ_(offset). That is, Implementation 2 implementsa complex value derived on a sample-by-sample basis to rotate the phaseon a sample-by-sample basis. By contrast, Implementation 3 onlymultiplies the input to the FFT by a fixed complex value on an OFDMsymbol-by-symbol basis.

In addition, by way of comparison, Implementation 1 utilizes 128 or128*3 complex-valued sequences for phase compensation for all availablecarrier frequency candidates. That is, since Implementation 1 utilizesone of 128 or 128*3 complex values per symbol for each carrierfrequency, Implementation 1 utilizes 128 or 128*3 complex-valuedsequences for phase compensation according to carrier frequencies. Bycontrast, Implementation 3 does not utilize 128 or 128*3 complex-valuedsequences for f_(frac) because Implementation 3 performs frequency shiftby f_(rrac) by changing the resource de-mapping position from the FFT,and only performs phase compensation for Δ_(offset) that is implementedthrough analog frequency downconversion. Therefore, Implementation 3utilizes only two complex-valued sequences corresponding to +/−5 kHz forphase compensation for Δ_(offset)=+/−5 kHz. In addition, since thenegative frequency corresponds to the opposite phase of a positivefrequency, Implementation 3 actually utilizes only one complex-valuedsequence for phase compensation. In some implementations, in the case offrequency ranges FR2 and FR1 (>3 GHz) or in the case of Δ_(offset)=0 infrequency range FR1 (<3 GHz), the function of downconversion from thecarrier frequency may be configured without any additional operation(e.g., frequency shift by Δ_(offset)) other than FFT and downconversion,as in the case of Implementation 2.

The implementations described above (Implementation 1, Implementation 2and Implementation 3) have been discussed for scenarios where phasecompensation for phase discontinuity is performed on the carrierfrequency f_(Tx) of the transmitting side and the carrier frequencyf_(Rx) of the receiving side. However, referring to Equation 16, above,the phase compensation ultimately corresponds to correcting the phasediscontinuity (i.e., phase mismatch) corresponding to f_(Tx)−f_(Rx).Since f_(Tx) and f_(Rx) correspond to positions of subcarriers (e.g.,integer multiples of the subcarrier spacing), phase compensation mayalso be performed in scenarios where Δ_(offset) is considered to bezero. For example, even if Δ_(offset) is not actually equal to 0, anassumption of Δ_(offset) being equal to 0 may be correspond to thescenario of phase correction/compensation for Δ_(offset) not beingperformed. As such, implementations of the present disclosure may alsobe applied to a case where Δ_(offset) is considered to be 0. IfΔ_(offset) is considered to be equal to 0, then phase compensation maybe performed only for the frequency magnitudes except Δ_(offset) at thecarrier frequency at the transmitting side and the receiving side. Inscenarios where Δ_(offset) is considered to be 0 in the phasecompensation term, the equality f_(Tx)=N_(Tx)*Δf holds, and thereforethe terms Ψ_(Tx,l)=−2πf_(Tx)(t_(start,l)+N_(CP,l) ^(μ)T_(c)) andΨ_(Rx,l)=2πf_(Rx)(t_(start,l)+N_(CP,l) ^(μ)T_(c)) corresponding to phasecompensation in the last line of Equation 16 may be expressed asfollows.Ψ_(Tx,l)=−2π(N _(Tx) Δf)·(t _(start,l) +N _(CP,l) ^(μ) T _(c))  Equation28Ψ_(Rx,l)=−2π(N _(Rx) Δf)·(t _(start,l) +N _(CP,l) ^(μ) T _(c))  Equation29

In Equation 28, the following equalities hold:f_(Tx)=N_(Tx)*Δf+Δ_(offset)=N_(int)*128Δf+N_(frac)*+Δ_(offset)=f_(base)+N_(frac)*Δf+Δ_(offset)=f_(base)+f_(frac)+Δ_(offset).Furthermore, in Equation 29, the following equalities hold:f_(Rx)=N_(Rx)*Δf+Δ_(offset)=N_(int)*128Δf+N_(frac)+Δf+Δ_(offset)=f_(base)+N_(frac)*Δf+Δ_(offset)=f_(base)+f_(frac)+Δ_(offset).The OFDM symbol signal generation/recovery according to the presentdisclosure, which is performed assuming Δ_(offset) be 0, may be appliedsimilarly to offset to Implementation 1, Implementation 2-1, andImplementation 2-2 described above, except that phase compensation forΔ_(offset) is not performed assuming that Δ_(offset) is 0.

In such scenarios, Implementation 1, Implementation 2-1, Implementation2-2, and Implementation 3 may be implemented as follows, correspondingto Implementation a1, Implementation a2-1, Implementation a2-2, andImplementation a3, respectively.

Implementation a1

Equation 19 relating to the transmitting side of Implementation 1 isgiven below again.

$\begin{matrix}{{x^{({p,\mu})}(t)} = {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot \{ {e^{{{j2\pi}{({k + k_{0} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot e^{{j\Psi}_{{Tx},l}}} \} \cdot e^{{j2\pi f}_{{Tx}^{t}}}}}} & {{Equation}\mspace{14mu} 30}\end{matrix}$

The phase compensation term of Implementation 1 is given byΨ_(Tx,l)=−2π(N_(frac)Δf+Δ_(offset))·(t_(start,l)+N_(CP,l) ^(μ)T_(c)),but the phase compensation term of Implementation a1, which does notconsider Δ_(offset), is given byΨ_(Tx,l)=−2π(N_(frac)Δf)·(t_(start,l)+N_(CP,l) ^(μ)T_(c)).

Among Equations 22 and 23 relating to the receiving side ofImplementation 1, Equation 23 is given below again.

$\begin{matrix}{{\overset{\sim}{a}}_{k,l}^{({p,\mu})} = {e^{{j\Psi}_{{Rx},l}} \cdot {\sum\limits_{i}\{ {( {{y^{({p,\mu})}(t)} \cdot e^{- {j2\pi f}_{{Rx}^{l}}} \cdot {\delta( {t - {i \cdot T_{c}}} )}} ) \cdot e^{{- {{j2\pi}{({k + k_{0,{Rx}} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}}{{\Delta f}{({{i \cdot T_{c}} - {({t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}})}}}} \}}}} & {{Equation}\mspace{14mu} 31}\end{matrix}$

The phase compensation term of Implementation 1 is given byΨ_(Rx,l)=2π(N_(frac)Δf+Δ_(offset))·(t_(start,l)+N_(CP,l) ^(μ)T_(c)), butthe phase compensation term of Implementation a1 is given byΨ_(Rx,l)=2π(N_(Rx)Δf)·(t_(start,l)+N_(CP,l) ^(μ)T_(c)) in scenarioswhere Δ_(offset) is not considered.

Phase compensation of Implementation 1 is performed using one of 128*3complex-valued sequences for a frequency band where Δ_(offset) may be −5kHz, 0 or +5 kHz, for example. By contrast, phase compensation ofImplementation a1 is performed using one of 128 complex valuedsequences, regardless of Δ_(offset). In Implementation a1, even if thefrequency shift for Δ_(offset) is performed by an analog free-runningOSC, phase compensation is not performed to correct phase mismatch thatmay occur due to Δ_(offset).

Implementation a2-1

FIGS. 10A and 10B are diagrams illustrating examples of Implementationa2-1 of the present disclosure. Specifically, FIG. 10A shows a part ofthe transmitting side structure according to Implementation a2-1, andFIG. 10B shows a part of the receiving side structure according toImplementation a2-1.

Considering that phase compensation is performed only for componentsother than Δ_(offset), Equation 24 relating to the transmitting side ofImplementation 2-1 may be changed to the following equation.

$\begin{matrix}\begin{matrix}{{x^{({p,\mu})}(t)} = {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ e^{{{j2\pi}{({k + k_{0} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \} \cdot} \\{e^{{{j2\pi}{({f_{base} + f_{frac}})}}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}} \cdot e^{{{j2\pi}\Delta}_{{offset}^{t}}}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ {e^{{{j2\pi}{({k + k_{0} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot} } \\{ e^{{j2\pi N}_{frac}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \} \cdot} \\{e^{{j2\pi f}_{base}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot e^{{{j2\pi}\Delta}_{{offset}^{t}}}} \\{= {\sum\limits_{k = 0}^{{N_{grid}^{{size},\mu}N_{sc}^{RB}} - 1}{a_{k,l}^{({p,\mu})} \cdot}}} \\{\{ e^{{{j2\pi}{({k + k_{0} + N_{frac} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}{{\Delta f}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \} \cdot} \\{e^{{{j2\pi}{({f_{base} + f_{offset}})}}^{t}}}\end{matrix} & {{Equation}\mspace{14mu} 32}\end{matrix}$

The first line of Equation 32 may be obtained by substitutingf_(base)+f_(frac)+Δ_(offset) for f_(Tx) in the first line of Equation24. As seen from Equation 32, the phase compensation of Implementationa2-1 may be performed by changing the mapping position of a resource(corresponding to the term k+k₀+N_(frac)−N_(grid) ^(size,μ)N_(sc)^(RB)/2) in the IFFT term

e^(j 2 π (k + k₀ + N_(frac) − N_(grid)^(size, μ)N_(sc)^(RB)/2)Δ f(t − t_(start, l) − N_(CP, l)^(μ)T_(c)))and performing upconversion (corresponding to the term e^(j2π(f) ^(base)^(+Δ) ^(offset) ⁾ ^(t) ) to a frequency (i.e., f_(base)+Δ_(offset))obtained by adding Δ_(offset) to f_(base), which is a frequency closestto f_(Tx) among the frequencies corresponding to integer multiples of128Δf (e.g., among frequencies less than or equal to f_(TX) or amongfrequencies greater than or equal to f_(Tx) or among frequencies on bothsides of f_(TX)). Consequently, if phase compensation for Δ_(offset) isnot performed, then upconversion to the carrier frequency may beperformed through the resource mapping shift for the IFFT andupconversion (using an analog OSC) as illustrated in FIG. 10A. InImplementation a2-2, even if the frequency shift for Δ_(offset) isperformed by the analog free-running OSC, the Δ_(offset) is consideredto be 0, namely, phase correction/compensation for Δ_(offset) is notperformed.

Analogously, that phase compensation is performed only for componentsconsidering other than Δ_(offset), Equation 25 relating to the receivingside in Implementation 2-1 may be modified to the following equation.

$\begin{matrix}\begin{matrix}{{\overset{\sim}{a}}_{k,l}^{({p,\mu})} = {\sum\limits_{i}\{ ( {{y^{({p,\mu})}(t)} \cdot e^{{- {{j2\pi}{({f_{base} + f_{frac}})}}}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}} \cdot}  }} \\{ {e^{- {{j2\pi}\Delta}_{{offset}^{t}}} \cdot {\quad{\delta( {t - {i \cdot T_{c}}} )}}} ) \cdot} \\ e^{{- {{j2\pi}{({k + k_{0,{Rx}} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}}{{\Delta f}{({{i \cdot T_{c}} - {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}})}}} \} \\{= {\sum\limits_{i}\{ ( {{y^{({p,\mu})}(t)} \cdot e^{- {{j2\pi f}_{base}{({t - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \cdot}  }} \\{ {e^{- {{j2\pi}\Delta}_{{offset}^{l}}} \cdot {\quad{\delta( {t - {i \cdot T_{c}}} )}}} ) \cdot} \\{e^{{- {{j2\pi}{({k + k_{0,{Rx}} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}}{{\Delta f}{({{i \cdot T_{c}} - {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}})}}} \cdot} \\ e^{- {{j2\pi N}{({{i \cdot T_{c}} - t_{{start},l} - {N_{{CP},l}^{\mu}T_{c}}})}}} \} \\{ {= {\sum\limits_{i}\{ ( {{y^{({p,\mu})}(t)} \cdot e^{- {{j2\pi}{({f_{base} + f_{frac}})}}^{l}} \cdot {\quad{\delta( {t - {i \cdot T_{c}}} )}}}  }} ) \cdot} \\ e^{{- {{j2\pi}{({k + k_{0,{Rx}} + N_{frac} - {N_{grid}^{{size},\mu}{N_{sc}^{RB}/2}}})}}}{{\Delta f}{({{i \cdot T_{c}} - {({t_{{start},l} + {N_{{CP},l}^{\mu}T_{c}}})}})}}} \}\end{matrix} & {{Equation}\mspace{14mu} 33}\end{matrix}$

The first line of Equation 33 may be obtained by substitutingf_(base)+f_(frac)+Δ_(offset) for f_(Rx) in the first line of Equation25. As can be seen from Equation 33, the phase compensation ofImplementation 2-1a includes changing the de-mapping position of aresource (corresponding to the term k+k_(0,Rx)+N_(frac)−N_(grid)^(size,μ)N_(sc) ^(RB)/2) in the FFT term

e^(−j 2 π (k + k_(0, Rx) + N_(frac) − N_(grid)^(size, μ)N_(sc)^(RB)/2)Δ f(t − t_(start, l) − N_(CP, l)^(μ)T_(c)))and downconversion (corresponding to the term e^(j2π(f) ^(base) ^(+Δ)^(offset) ^()t)) by a frequency (i.e., f_(base)+Δ_(offset)) obtained byadding Δ_(offset) to f_(base), which is a frequency closest to f_(Rx)among the frequencies corresponding to integer multiples of 128Δf (amongfrequencies less than or equal to f_(Rx) or among frequencies greaterthan or equal to f_(Rx) or among frequencies on both sides of f_(Rx)).Consequently, if phase compensation for Δ_(offset) is not performed,then the phase compensation function may be performed through theresource de-mapping shift for the FFT and downconversion (using ananalog OSC) as illustrated in FIG. 10B.

Implementation a2-2

FIGS. 11A and 11B are diagrams illustrating examples of Implementationa2-2 of the present disclosure. Specifically, FIG. 11A shows a part ofthe transmitting side structure according to Implementation a2-2, andFIG. 11B shows a part of the receiving side structure according toImplementation a2-2. In FIGS. 11A and 11B, the term t_(l) denotes thestart position of the signal part of an OFDM symbol l in the timedomain, and may be expressed as t_(l)=t_(start,l)+N_(CP,l) ^(μ)T_(c).

Referring first to FIG. 11A, showing a part of the transmitting inImplementation a2-2, the offset Δ_(offset) is considered to be 0, i.e.,phase correction/compensation for Δ_(offset) is not performed. That is,even if frequency shift is performed by f_(base)+Δ_(offset) using ananalog free-running OSC, phase correction/compensation for Δ_(offset) isnot performed.

By way of comparison, in Implementation a2-1 described above, at thetransmitter side, the frequency corresponding to f_(base) and Δ_(offset)is used for the upconversion process by the OSC of the RF stage (thefree-running OSC) at the transmitting side, and the portioncorresponding to f_(frac) is used in the process of determining theresource mapping position with respect to the IFFT. However, in somescenarios of Implementation a2-1, described above, the degree ofasymmetry of a signal output in the baseband may be increased withrespect to DC depending on the value of f_(frac). This may limit theefficiency of the spectrum by filtering after the IFFT output in thetransmitter (or before the input of the FFT in the receiver).

Therefore, to address such problems, in some scenarios it may bedesirable to modify the transmitting side operations of Implementationa2-1, described above, for implementing the function of frequencyupconversion to f_(frac) (i.e., frequency up-shift by f_(frac)) andphase reset by adjusting the position of the resource mapping of IFFT.

Accordingly, to this end, in Implementation a2-2, at the transmitterside, the function of frequency shift by f_(frac) and phase reset may berealized by a digital OSC. As such, as illustrated in FIG. 11A, in orderto perform frequency upconversion by f_(Tx), Implementation a2-2 uses afrequency corresponding to f_(base) and Δ_(offset) in f_(Tx) forupconversion by the free-running OSC at the RF stage, and uses f_(frac)in performing frequency upconversion on an OFDM symbol-by-symbol basisand resetting the phase (e.g. to have the zero phase at the end time ofthe cyclic prefix CP) through the digital OSC.

Referring next to FIG. 11B, showing a part of the receiving side inImplementation a2-2, the offset Δ_(offset) is considered to be 0,namely, phase correction/compensation for Δ_(offset) is not performed.As such, even if frequency shift by f_(base)+Δ_(offset) is performed bythe analog free-running OSC, phase correction/compensation forΔ_(offset) is not performed.

By way of comparison, in Implementation a2-1 described above, at thereceiving side, the frequency corresponding to f_(base) and Δ_(offset)is used for the downconversion process by the OSC of the RF stage (thefree-running OSC) at the receiving side, and the portion correspondingto f_(frac) is used in the process of determining the resourcede-mapping position with respect to the FFT. However, in some scenariosof Implementation a2-1, described above, the degree of asymmetry of asignal (i.e., the RF output) output in downconversion may be increasedwith respect to DC depending on the value of f_(frac). This may limitthe efficiency of the spectrum by filtering after downconversion outputin the receiver (or before the input of the FFT in the receiver).

Therefore, to address such problems, in some scenarios it may bedesirable to modify the receiving side operations of Implementationa2-1, described above, for implementing the function of frequencydownconversion (i.e., frequency down-shift) and phase resetcorresponding to f_(frac) by adjusting the position of the resourcede-mapping of the FFT.

Accordingly, to this end, in Implementation a2-2, at the receiving side,the function of frequency shift by f_(frac) and phase reset may berealized by a digital OSC. As such, as illustrated in FIG. 11B, in orderto perform frequency downconversion by f_(Rx), Implementation a2-2 usesa frequency corresponding to f_(base) and Δ_(offset) in f_(Tx) fordownconversion by the free-running OSC (the analog OSC) at the RF stage,and uses f_(frac) in performing frequency downconversion on an OFDMsymbol-by-symbol basis and resetting the phase (e.g. to have the zerophase at the end time of the cyclic prefix CP) through the digital OSC.

To recap, Implementation 1, Implementation 2, and Implementation 3 arebriefly summarized below.

Implementation 1 predetermines 128 compensation-valued sequences (or128*3 compensation-valued sequences if there are three values ofΔ_(offset)) for phase compensation, and performs phase compensation onthe corresponding OFDM symbol using one compensation-valued sequence forthe corresponding carrier frequency among 128 or 128*3 predeterminedcompensation-valued sequences.

In Implementation 2, an integer multiple of 128Δf is used as the basecarrier frequency f_(base) in both Implementation 2-1 and Implementation2-2, and frequency up/down-shift is performed by f_(base) by thefree-running OSC at the RF stage. In Implementation 2, the transmittingside compensates for the frequency difference between f_(TX) andf_(base) by resource mapping for IFFT (Implementation 2-1), orcompensates for the difference by frequency shift using the digital OSCafter IFFT (Implementation 2-2). The receiving side of Implementation 2compensates for the frequency difference between f_(Tx) and f_(base) byresource de-mapping for FFT (Implementation 2-1), or compensates for thedifference by frequency shift using the digital OSC before FFT(Implementation 2-2).

Implementation 3 is similar to Implementation 2 except thatImplementation 3 performs phase compensation for Δ_(offset) on an OFDMsymbol-by-symbol basis, unlike Implementation 2, which resets the phaseon a sample-by-sample basis for frequency shift by Δ_(offset).

FIG. 12 is a diagram illustrating another use example of the presentdisclosure.

As described above, in order to prevent phase mismatch between OFDMsymbols or to facilitate phase compensation, a frequency correspondingto an integer multiple of 128Δf is used by an analog OSC in the processof OFDM symbol signal generation or OFDM symbol signal recovery. Forexample, when a numerology of a frequency band supporting a plurality ofnumerologies is changed, for example, when the subcarrier spacing (SCS)of the frequency band is changed from 30 kHz to 15 kHz, or vice versa,frequencies which are integer multiples of 128Δf may be mismatched. Forexample, suppose that a frequency corresponding to an integer multipleof 128Δf closest to f_(Tx) is f_(base,1) when Δf=30 kHz (wheref_(base,1)=N_(int,1)*128Δf for Δf=30 kHz). If the SCS changes from Δf=30kHz to Δf=15 kHz in the same frequency band as f_(Tx), the frequencyf_(base,0) corresponding to an integer multiple of 128Δf closest to theupconversion frequency f_(Tx) when Δf=15 kHz (wheref_(base,0)=N_(int,0)*128Δf for Δf=15 kHz) may differ from f_(base,1) byΔf_(base). Alternatively, the upconversion/downconversion frequency maybe varied by Δf_(base) according to change in SCS. In this case, thetransmitter and the receiver of the present disclosure may compensatefor Δf_(base) using a digital OSC or IFFT/FFT resourcemapping/de-mapping according to Implementation 2 of the presentdisclosure described above, rather than performing RF retuning.

FIGS. 13A and 13B illustrate examples of a transmitter structure and areceiver structure according to the present disclosure. The transmittingside structure and the receiving side structure of the presentdisclosure described based on the basic structure of FIGS. 13A and 13B.

Referring to FIG. 13A, the transmitter generates symbols (hereinafterreferred to as information symbols) for bit sequences input accordingto, for example, a signal generation technique defined in the standard.The transmitter performs, at the input side of the IFFT, appropriateresource mapping (i.e., subcarrier mapping) of the generated informationsymbols in accordance with the band in which transmission is to beperformed, and performs IFFT, which converts a frequency-domain signalinto a time-domain signal, on the resource-mapped information symbols.The transmitter inserts a cyclic prefix (CP), which is configured formitigation/avoidance of interference between OFDM symbols, into the IFFToutput. For reference, while FIGS. 5A to 11B illustrate that IFFT/FFTinclude the resource mapping/de-mapping function and the CPattach/detach function, the CP attach/detach function may be implementedseparately from the FFT/IFFT as shown in FIG. 13. For a signal generatedthrough IFFT and CP attachment, the transmitter may perform filtering orwindowing to satisfy spectral characteristics before performingupconversion to a carrier frequency. However, filtering or windowing maynot be a function that must be implemented depending on thecharacteristics of an RF device. In order to transmit a signal generatedvia IFFT and CP attachment or a signal generated via IFFT and CPattachment (and filtering/windowing) using a predefined carrierfrequency, the transmitter performs unconversion of the signal to thepredefined carrier frequency. In general, the upconversion is performedusing a digital-to-analog converter (DAC) for converting a digitalsignal into an analog signal, an oscillator and a PLL (Phase-LockedLoop) for generating a carrier frequency, a mixer for shifting abaseband signal to a desired carrier frequency, and the like.Thereafter, the transmitter transmits the frequency up-converted signalto the outside through an analog filter, an amplifier, and an antenna.

Since the signals input to the digital-to-analog converter in thetransmitter are digital signals and the signals output from thedigital-to-analog converter are analog signals, the transmitter modulesused for signal processing before the digital-to-analog converter may bedigital modules, and the transmitter modules used for signal processingafter the digital-to-analog converter may be analog modules.

The receiver performs an operation corresponding to the inverse processof the transmitter. In the receiver operations, a signal that thetransmitter transmits is received by the receiver through the antenna,amplifier and analog filter of the receiver. Referring to FIG. 13B, thereceiver performs downconversion on the received signal. In general, thedownconversion is performed using an analog-to-digital converter (ADC)for converting an analog signal into a digital signal, an oscillator andPLL for generating a carrier frequency, a mixer for shifting a signalreceived through the carrier frequency to a band signal, and the like.The receiver may filter the signal transmitted through the basebandaccording to the spectral characteristics. Filtering may not beimplemented depending on the characteristics of the RF devices. Thereceiver detaches the cyclic prefix (CP) from the (filtered orunfiltered) baseband signal according to the pre-measured timinginformation, and converts the CP-detached signal into a frequency-domainsignal through FFT for converting a time-domain signal into afrequency-domain signal. The FFT function includes a resource de-mappingfunction for deriving only a signal transmitted to the receiver fromamong the entire frequency-domain signals. The receiver recovers thesignal transmitted by the transmitter from a resource de-mapped signalthrough a symbol recovery process for compensating for a distorted parton the channel, performs a decoding process for a specific signalgeneration technique, for example, a signal generation technique definedby the communication standard, and then obtains the final signal (bitsequence). Both the process of compensating for the distorted part onthe channel and the decoding process correspond to the symbol recoveryprocess.

Since the signals input to the analog-to-digital converter in thereceiver are analog signals and the signals output from theanalog-to-digital converter are digital signals, the receiver modulesused for signal processing before the analog-to-digital converter may beanalog modules, and the receiver modules used for signal processingafter the analog-to-digital converter may be digital modules.

Although not shown in FIGS. 13A and 13B, the transmitter and thereceiver may include a digital oscillator configured to performoperations according to the present disclosure.

FIG. 14 is a block diagram illustrating examples of elements of atransmitting device 10 and a receiving device 20 for implementing thepresent disclosure.

The transmitting device 10 and the receiving device 20 respectivelyinclude radio frequency (RF) units 13 and 23 capable of transmitting andreceiving radio signals carrying information, data, signals, and/ormessages, memories 12 and 22 for storing information related tocommunication in a wireless communication system, and processors 11 and21 operationally connected to elements such as the RF units 13 and 23and the memories 12 and 22 to control the elements and configured tocontrol the memories 12 and 22 and/or the RF units 13 and 23 so that acorresponding device may perform at least one of the above-describedimplementations of the present disclosure.

The memories 12 and 22 may store programs for processing and controllingthe processors 11 and 21 and may temporarily store input/outputinformation. The memories 12 and 22 may be used as buffers.

The processors 11 and 21 generally control the overall operation ofvarious modules in the transmitting device and the receiving device.Especially, the processors 11 and 21 may perform various controlfunctions to implement the present disclosure. The processors 11 and 21may be referred to as controllers, microcontrollers, microprocessors, ormicrocomputers. The processors 11 and 21 may be implemented by hardware,firmware, software, or a combination thereof. In a hardwareconfiguration, application specific integrated circuits (ASICs), digitalsignal processors (DSPs), digital signal processing devices (DSPDs),programmable logic devices (PLDs), or field programmable gate arrays(FPGAs) may be included in the processors 11 and 21. In someimplementations, if the present disclosure is implemented using firmwareor software, the firmware or software may be configured to includemodules, procedures, functions, etc. performing the functions oroperations of the present disclosure. Firmware or software configured toperform the present disclosure may be included in the processors 11 and21 or stored in the memories 12 and 22 so as to be driven by theprocessors 11 and 21.

In some scenarios of the present disclosure, functions, procedures,and/or methods disclosed in the present disclosure may be implemented bya processing chip (also referred to as a processing device). Theprocessing chip may be a system on chip (SoC). The processing chip mayinclude the processor 11 and/or 21 and the memory 12 and/or 22, and maybe mounted on, installed on, or connected to the transmitting device 10or the receiving device 20. The processing chip may be configured toperform or control any one of the methods and/or processes describedherein and/or to cause such methods and/or processes to be performed bya communication device which the processing chip is mounted on,installed on, or connected to. The memories 12 and 22 in the processingchip may be configured to store software codes including instructionsthat, when executed by the processor, causes the processors 11 and 21 toperform some or all of functions, methods or processes discussed in thepresent disclosure. The memories 12 and 22 in the processing chip maystore or buffer information, data or signals generated by the processorof the processing chip or information, data or signals recovered orobtained by the processors 11 and 21 of the processing chip. One or moreprocesses involving transmission or reception of the information, dataor signals may be performed by the processors 11 and 21 of theprocessing chip or under control of the processors 11 and 21 of theprocessing chip. For example, the RF units 13 and 23 operably connectedor coupled to the processing chip may transmit or receive signalscontaining the information or data under the control of the processor 11and 21 of the processing chip.

The processor 11 mounted on, installed on, or connected to thetransmitting device 10 performs predetermined coding and modulation fora signal and/or data scheduled to be transmitted to the outside by theprocessor 11 or a scheduler connected with the processor 11, and thentransfers the coded and modulated data to the RF unit 13. For example,the processor 11 converts a data stream to be transmitted into K layersthrough demultiplexing, channel coding, scrambling, and modulation. Thecoded data stream is also referred to as a codeword and is equivalent toa transport block which is a data block provided by a MAC layer. Onetransport block (TB) is coded into one codeword and each codeword istransmitted to the receiving device in the form of one or more layers.The processor 11 may determine or generate symbols (hereinafter referredto as information symbols) for bit sequences input according to, forexample, a signal generation technique defined in the standard. Theprocessor 11 may determine a carrier frequency for transmission of radiosignals. The processor 11 may determine the frequency f_(base) used forfrequency upconversion. The processor 11 may determine the frequencyf_(base) among frequencies which are integer multiples of

${\frac{N_{sample}}{gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}*{\Delta f}},$based on the carrier frequency. For frequency up-conversion, the RF unit13 may include an oscillator. The RF unit 13 may include N_(t) (whereN_(t) is a positive integer) transmit antennas. The RF unit 13 mayperform frequency upconversion by an oscillator according to the presentdisclosure under control of the processor 11 to generate an OFDM symbolsignal. For example, in the case of Implementation 2, the processor 11may control the oscillator (i.e., the analog oscillator) of the RF unit13 so as to perform upconversion using a frequency that is an integermultiple of

$\frac{N_{sample}}{gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}*{{\Delta f}.}$

A signal processing process of the receiving device 20 is the reverse ofthe signal processing process at the transmitting device 10. Undercontrol of the processor 21, the RF unit 23 of the receiving device 20receives radio signals transmitted by the transmitting device 10. The RFunit 23 may include N_(r) receive antennas, and the RF unit 23 mayperform frequency downconversion of each signal received through thereceive antennas by an oscillator according to the present disclosureunder control of the processor 21 to recover the baseband signal. Theprocessor 21 may determine a carrier frequency for reception of radiosignals. The processor 21 may determine a frequency f_(base) used forfrequency downconversion. The processor 21 may determine the frequencyf_(base) among frequencies which are integer multiples of

${\frac{N_{sample}}{gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}*{\Delta f}},$based on the carrier frequency. For example, in the case ofImplementation 2, the processor 21 may control the oscillator (i.e., theanalog oscillator) of the RF unit 23 so as to perform down conversionusing a frequency that is an integer multiple of

$\frac{N_{sample}}{gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}*{{\Delta f}.}$The RF unit 23 may include an oscillator for frequency downconversion.The processor 21 may perform decoding and demodulation of the radiosignal received through the receive antenna to recover data that thetransmitting device 10 originally intended to transmit.

The RF units 13 and 23 include one or more antennas. An antenna performsa function for transmitting signals processed by the RF units 13 and 23to the exterior or receiving radio signals from the exterior to transferthe radio signals to the RF units 13 and 23. The antenna may also becalled an antenna port. Each antenna may correspond to one physicalantenna or may be configured by a combination of more than one physicalantenna element. The signal transmitted from each antenna cannot befurther deconstructed by the receiving device 20. An RS transmittedthrough a corresponding antenna defines an antenna from the view pointof the receiving device 20 and enables the receiving device 20 to derivechannel estimation for the antenna, irrespective of whether the channelrepresents a single radio channel from one physical antenna or acomposite channel from a plurality of physical antenna elementsincluding the antenna. That is, an antenna is defined such that achannel carrying a symbol of the antenna can be obtained from a channelcarrying another symbol of the same antenna. An RF unit supporting aMIMO function of transmitting and receiving data using a plurality ofantennas may be connected to two or more antennas.

In the present disclosure, a user equipment (UE), that is, a terminaloperates as the transmitting device 10 on the uplink and as thereceiving device 20 on the downlink. In the present disclosure, the basestation operates as the receiving device 20 on the uplink and as thetransmitting device 10 on the downlink.

The processor 11 mounted on, installed on, or connected to thetransmitting device 10 may be configured to perform processes accordingto the present disclosure on a signal to be transmitted, and may controlthe modules (see FIG. 13A) of the transmitter so as to performoperations according to the present disclosure on the signal to betransmitted or the signal processed. For example, for a frequency shiftcorresponding to the difference between f₀ and f_(base), the processor11 may control the IFFT so as to up-shift the resource mapping positionof the signal to be transmitted for the IFFT by N_(frac). As anotherexample, the processor 11 may control the digital oscillator to performfrequency up-shift by the difference between f₀ and f_(base). As anotherexample, the processor 11 may control the digital oscillator to resetthe phase to a certain value at the end of the cyclic prefix (CP) partof the OFDM symbol, i.e., at the beginning of the signal part of theOFDM symbol. The processor 11 may be configured to use a frequencyclosest to f₀ among frequencies that are integer multiples of

$\frac{N_{sample}}{gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}*{\Delta f}$as f_(base).

The processor 21 at the receiving device 10 may control the modules (seeFIG. 13B) of the receiver to perform the operations according to thepresent disclosure on the received signal, and be configured to performprocesses according to the present disclosure on the signals from the RFunit 23. For example, for a frequency shift corresponding to thedifference between f₀ and f_(base), the processor 21 may control the FFTso as to down-shift the resource de-mapping position from the FFT forthe received signal by N_(frac). As another example, the processor 21may control the digital oscillator so as to perform frequency down-shiftby the difference between f₀ and f_(base). As another example, theprocessor 21 may control the digital oscillator to reset the phase to acertain value at the end of the cyclic prefix (CP) part of the OFDMsymbol, i.e., at the beginning of the signal part of the OFDM symbol.The processor 21 may be configured to use a frequency closest to f₀among frequencies that are integer multiples of

$\frac{N_{sample}}{\gcd\{ {N_{{CP},1},N_{{CP},2},\ldots\mspace{14mu},N_{sample}} \}}*{\Delta f}$as f_(base).

The transmitting device 10 may be configured to include FIG. 13A. Thereceiving device 20 may be configured to include FIG. 13B. In theimplementations of the present disclosure described above, upconversionand downconversion by the free-running oscillator may be provided in theRF units 13, 23, and the other operations of the present disclosure(e.g., baseband signal generation, IFFT/FFT, resourcemapping/de-mapping, cyclic prefix (CP) attachment/detachment, filtering,symbol recovery) may be performed by the processors 11, 21 or undercontrol of the processors 11, 21.

While the transmitting device 10 and the receiving device 20 areseparately shown in FIG. 14, the processor 11, the memory 12 and the RFunit 13 in the transmitting device 10 may also be configured to performthe operations of the receiving device 20, and the processor 21, thememory 22 and the RF unit 23 in the receiving device 20 may also beconfigured to perform the operations of the transmitting device 10. Apart of the transmitter illustrated in FIG. 13A and a part of thereceiver illustrated in FIG. 13B may be implemented as a transceiver.Alternatively, the term “transceiver” may be used to refer to the RFunit 13 of the transmitting device 10 or the RF unit 23 of the receivingdevice 20. A part of the transmitter illustrated in FIG. 13A and a partof the receiver illustrated in FIG. 13B may be implemented with theprocessors 11, 21.

What is claimed are:
 1. A method of transmitting, by a transmittingdevice, an orthogonal frequency division multiplexing (OFDM) signal in awireless communication system, the method comprising: generating, by adigital module of the transmitting device, a frequency-shifted OFDMbaseband signal by performing frequency up-shift of a first signal by adifference between a carrier frequency f₀ and a first frequencyf_(base), wherein the first frequency f_(base) is, among frequenciescorresponding to integer multiples of 128Δf, closest to the carrierfrequency f₀, and wherein Δf is an OFDM subcarrier spacing;up-converting, by an analog oscillator of the transmitting device, thefrequency-shifted OFDM baseband signal by the first frequency f_(base)to generate an OFDM symbol signal at the carrier frequency f₀; andtransmitting, by a transmitter of the transmitting device, the OFDMsymbol signal at the carrier frequency f₀.
 2. The method according toclaim 1, wherein the digital module is configured to implement aninverse fast Fourier transform (IFFT) on the first signal.
 3. The methodaccording to claim 2, wherein performing the frequency up-shift of thefirst signal by the difference between f₀ and f_(base) comprises:up-shifting, by N_(frac), a resource mapping for the first signal thatis input to the IFFT, where N_(frac) is an integer satisfyingf₀−f_(base)=N_(frac)*Δf.
 4. The method according to claim 1, wherein thedigital module comprises a digital oscillator, and wherein performingthe frequency up-shift of the first signal by the difference between f₀and f_(base) is performed by the digital oscillator.
 5. The methodaccording to claim 4, further comprising: resetting, by the digitaloscillator and before transmitting the OFDM symbol signal, a phase ofthe OFDM symbol signal to a predetermined value at an end of a cyclicprefix of the OFDM symbol signal.
 6. A method of receiving, by areceiving device, an orthogonal frequency division multiplexing (OFDM)signal in a wireless communication system, the method comprising:receiving an OFDM symbol signal at a carrier frequency f₀;down-converting, by an analog oscillator of the receiving device, theOFDM symbol signal by a first frequency f_(base) to generate adown-converted OFDM symbol signal; and generating, by a digital moduleof the receiving device, an OFDM baseband signal by performing frequencydown-shift of the down-converted OFDM symbol signal by a differencebetween the carrier frequency f₀ and f_(base), wherein the firstfrequency f_(base) is, among frequencies corresponding to integermultiples of 128Δf, closest to the carrier frequency f₀, and wherein Δfis an OFDM subcarrier spacing.
 7. The method according to claim 6,wherein the digital module is configured to implement a fast Fouriertransformer (FFT) on the down-converted OFDM symbol signal.
 8. Themethod according to claim 7, wherein performing the frequency down-shiftof the down-converted OFDM symbol signal by the difference between f₀and f_(base) comprises: down-shifting, by N_(frac), a resourcede-mapping from the FFT for the down-converted OFDM symbol signal, whereN_(frac) is an integer satisfying f₀−f_(base)=N_(frac)*Δf.
 9. The methodaccording to claim 6, wherein the digital module comprises a digitaloscillator, and wherein performing the frequency down-shift of thedown-converted OFDM symbol signal by the difference between f₀ andf_(base) is performed by the digital oscillator.
 10. The methodaccording to claim 9, further comprising: resetting, by the digitaloscillator, a phase of the down-converted OFDM symbol signal to apredetermined value at an end of a cyclic prefix of the down-convertedOFDM symbol signal.
 11. A transmitting device for transmitting anorthogonal frequency division multiplexing (OFDM) signal in a wirelesscommunication system, the transmitting device comprising: a digitalmodule; an analog oscillator; at least one antenna; at least oneprocessor; and at least one computer memory that is operably connectableto the at least one processor and that has stored thereon instructionswhich, when executed, cause the at least one processor to performoperations comprising: generating, by the digital module, afrequency-shifted OFDM baseband signal by performing frequency up-shiftof a first signal by a difference between a carrier frequency f₀ and afirst frequency f_(base), wherein the first frequency f_(base) is, amongfrequencies corresponding to integer multiples of 128Δf, closest to thecarrier frequency f₀, and wherein Δf is an OFDM subcarrier spacing;up-converting, by the analog oscillator, the frequency-shifted OFDMbaseband signal by the first frequency f_(base) to generate an OFDMsymbol signal at the carrier frequency f₀; and transmitting, using theat least one antenna, the OFDM symbol signal at the carrier frequencyf₀.
 12. The transmitting device according to claim 11, wherein thedigital module is configured to implement an inverse fast Fouriertransform (IFFT) on the first signal.
 13. The transmitting deviceaccording to claim 12, wherein performing the frequency up-shift of thefirst signal by the difference between f₀ and f_(base) comprises:up-shifting, by N_(frac), a resource mapping for the first signal thatis input to the IFFT, where N_(frac) is an integer satisfyingf₀−f_(base)=N_(frac)*Δf.
 14. The transmitting device according to claim11, wherein the digital module comprises a digital oscillator, andwherein performing the frequency up-shift of the first signal by thedifference between f₀ and f_(base) is performed by the digitaloscillator.
 15. The transmitting device according to claim 14, whereinthe operations further comprise: resetting, by the digital oscillatorand before transmitting the OFDM symbol signal, a phase of the OFDMsymbol signal to a predetermined value at an end of a cyclic prefix ofthe OFDM symbol signal.
 16. A receiving device for receiving anorthogonal frequency division multiplexing (OFDM) signal in a wirelesscommunication system, the receiving device comprising: at least oneantenna; an analog oscillator; a digital module; at least one processor;and at least one computer memory that is operably connectable to the atleast one processor and that has stored thereon instructions which, whenexecuted, cause the at least one processor to perform operationscomprising: receiving, using the at least one antenna, an OFDM symbolsignal at a carrier frequency f₀; down-converting, by the analogoscillator, the OFDM symbol signal by a first frequency f_(base) togenerate a down-converted OFDM symbol signal; and generating, by thedigital module, an OFDM baseband signal by performing frequencydown-shift of the down-converted OFDM symbol signal by a differencebetween the carrier frequency f₀ and f_(base), wherein the firstfrequency f_(base) is, among frequencies corresponding to integermultiples of 128Δf, closest to the carrier frequency f₀, and wherein Δfis an OFDM subcarrier spacing.
 17. The receiving device according toclaim 16, wherein the digital module is configured to implement a fastFourier transformer (FFT) on the down-converted OFDM symbol signal. 18.The receiving device according to claim 17, wherein performing thefrequency down-shift of the down-converted OFDM symbol signal by thedifference between f₀ and f_(base) comprises: down-shifting, byN_(frac), a resource de-mapping from the FFT for the down-converted OFDMsymbol signal, where N_(frac) is an integer satisfyingf₀−f_(base)=N_(frac)*Δf.
 19. The receiving device according to claim 16,wherein the digital module comprises a digital oscillator, and whereinperforming the frequency down-shift of the down-converted OFDM symbolsignal by the difference between f₀ and f_(base) is performed by thedigital oscillator.
 20. The receiving device according to claim 19,wherein the operations further comprise: resetting, by the digitaloscillator, a phase of the down-converted OFDM symbol signal to apredetermined value at an end of a cyclic prefix of the down-convertedOFDM symbol signal.